Radar apparatus

ABSTRACT

A radar transmitter includes: a plurality of transmission antennas that transmit a transmission signal; and a transmission circuit that applies a phase rotation amount corresponding to a Doppler shift amount and a code sequence to the transmission signal to perform multiplexing transmission of the transmission signal from the plurality of transmission antennas. Each of the plurality of transmission antennas is associated with each of a plurality of combinations of the Doppler shift amount and the code sequence. Each of the plurality of combination is different at least one of the Doppler shift amount and the code sequence, and the Doppler shift amounts of those of the plurality of combinations which are associated respectively with at least two transmission antennas of the plurality of transmission antennas are the same Doppler shift amount, the at least two transmission antennas being a first sub-array antenna.

TECHNICAL FIELD

The present disclosure relates to a radar apparatus.

BACKGROUND ART

Recently, studies have been developed on radar apparatuses that use aradar transmission signal of a short wavelength including a microwave ora millimeter wave that can achieve high resolution. Further, it has beendemanded to develop a radar apparatus which senses small objects such aspedestrians in addition to vehicles in a wide-angle range (e.g.,referred to as “wide-angle radar apparatus”) in order to improve theoutdoor safety.

Examples of the configuration of the radar apparatus having a wide-anglesensing range include a configuration using a technique of receiving areflected wave from a target by an array antenna composed of a pluralityof antennas (or also referred to as antenna elements), and estimatingthe direction of arrival of the reflected wave (or referred to as theangle of arrival) using a signal processing algorithm based on receivedphase differences with respect to element spacings (antenna spacings)(Direction of Arrival (DOA) estimation). Examples of the DOA estimationinclude a Fourier method, and, a Capon method, Multiple SignalClassification (MUSIC), and Estimation of Signal Parameters viaRotational Invariance Techniques (ESPRIT) that are methods achievinghigher resolution.

In addition, a radar apparatus has been proposed which, for example,includes a plurality of antennas (array antenna) at a transmitter sidein addition to at a receiver side, and is configured to perform beamscanning through signal processing using the transmission and receptionarray antennas (also referred to as Multiple Input Multiple Output(MIMO) radar) (e.g., see Non-Patent Literature (hereinafter referred toas “NPL”) 1).

CITATION LIST Patent Literature PTL 1

-   Japanese Unexamined Patent Application Publication (Translation of    PCT Application) No. 2011-526371

PTL 2

-   Japanese Patent Application Laid-Open No. 2014-119344

PTL 3

-   U.S. Pat. No. 9,541,638

Non Patent Literature NPL 1

-   J. Li, and P. Stoica, “MIMO Radar with Colocated Antennas,” Signal    Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007

NPL 2

-   M. Kronauge, H. Rohling, “Fast two-dimensional CFAR procedure,” IEEE    Trans. Aerosp. Electron. Syst., 2013, 49, (3), pp. 1817-1823

NPL 3

-   Direction-of-arrival estimation using signal subspace modeling    Cadzow, J. A.; Aerospace and Electronic Systems, IEEE Transactions    on Volume: 28, Issue: 1 Publication Year: 1992, Page(s): 64-79

SUMMARY OF INVENTION

However, methods for a radar apparatus (e.g., MIMO radar) to sense atarget object (or a target) have not been comprehensively studied.

One non-limiting and exemplary embodiment of the present disclosurefacilitates providing a radar apparatus with an enhanced sensingaccuracy for sensing a target object.

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes: a plurality of transmission antennas that transmita transmission signal; and a transmission circuit that applies a phaserotation amount corresponding to a Doppler shift amount and a codesequence to the transmission signal to perform multiplexing transmissionof the transmission signal from the plurality of transmission antennas,in which each of the plurality of transmission antennas is associatedwith a combination of the Doppler shift amount and the code sequencesuch that at least one of the Doppler shift amount and the code sequenceis different between a plurality of the combinations, and the Dopplershift amounts of those of the plurality of combinations which areassociated respectively with at least two adjacent transmission antennasof the plurality of transmission antennas are the same Doppler shiftamount, the at least two adjacent transmission antennas being a firstsub-array antenna.

Note that these generic or specific exemplary embodiments may beachieved by a system, an apparatus, a method, an integrated circuit, acomputer program, or a recoding medium, and also by any combination ofthe system, the apparatus, the method, the integrated circuit, thecomputer program, and the recoding medium.

According to an exemplary embodiment of the present disclosure, thetarget-object sensing accuracy of a radar apparatus can be improved.

Additional benefits and advantages of the disclosed exemplaryembodiments will become apparent from the specification and drawings.The benefits and/or advantages may be individually obtained by thevarious embodiments and features of the specification and drawings,which need not all be provided in order to obtain one or more of suchbenefits and/or advantages.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram illustrating an example configuration of aradar apparatus according to Embodiment 1;

FIG. 2 illustrates an example of a transmission signal in a case where achirp pulse is used;

FIG. 3 illustrates examples of assignment of Doppler shift amounts andorthogonal codes according to Embodiment 1;

FIG. 4 illustrates examples of the assignment of Doppler shift amountsand orthogonal codes according to Embodiment 1;

FIG. 5 illustrates examples of the assignment of Doppler shift amountsand orthogonal codes according to Embodiment 1;

FIG. 6 illustrates examples of the assignment of Doppler shift amountsand orthogonal codes according to Embodiment 1;

FIG. 7 illustrates examples of the assignment of Doppler shift amountsand orthogonal codes according to Embodiment 1;

FIG. 8 illustrates examples of the assignment of Doppler shift amountsand orthogonal codes according to Embodiment 1;

FIG. 9 illustrates examples of the assignment of Doppler shift amountsand orthogonal codes according to Embodiment 1;

FIG. 10 illustrates examples of the assignment of Doppler shift amountsand orthogonal codes according to Embodiment 1;

FIG. 11 illustrates examples of the assignment of Doppler shift amountsand orthogonal codes according to Embodiment 1;

FIG. 12 illustrates an arrangement example of transmission antennasaccording to Embodiment 1;

FIG. 13 illustrates an arrangement example of the transmission antennasaccording to Embodiment 1;

FIG. 14 illustrates an example of transmission signals and receptionsignals in a case where a chirp pulse is used;

FIG. 15 illustrates an example of Doppler domain compression processing;

FIG. 16 illustrates an example of Doppler aliasing judgement;

FIG. 17 illustrates an arrangement example of the transmission antennasaccording to Embodiment 1;

FIG. 18 illustrates an arrangement example of antennas;

FIG. 19 illustrates an arrangement example of antennas according toVariation 2 of Embodiment 1;

FIG. 20 illustrates an arrangement example of antennas according toVariation 3 of Embodiment 1;

FIG. 21 is a block diagram illustrating an example configuration of aradar apparatus according to Embodiment 2;

FIG. 22 is a block diagram illustrating another example configuration ofthe radar apparatus according to Embodiment 2;

FIG. 23 is a block diagram illustrating another example configuration ofthe radar apparatus according to Embodiment 1;

FIG. 24 is a block diagram illustrating still another exampleconfiguration of the radar apparatus according to Embodiment 2;

FIG. 25 illustrates an arrangement example of transmission antennasaccording to Embodiment 2;

FIG. 26 illustrates an arrangement example of the transmission antennasaccording to Embodiment 2;

FIG. 27 illustrates an example of transmission antennas of a sub-arrayconfiguration;

FIG. 28 illustrates an arrangement example of virtual reception antennasaccording to Embodiment 2;

FIG. 29 illustrates an arrangement example of the virtual receptionantennas; and

FIG. 30 illustrates one example of a direction estimation result

DESCRIPTION OF EMBODIMENTS

A MIMO radar transmits, from a plurality of transmission antennas (alsoreferred to as “transmission array antenna”), signals (radartransmission waves) that are time-division, frequency-division, orcode-division multiplexed, for example. The MIMO radar then receivessignals (radar reflected waves) reflected, for example, by an objectaround the radar using a plurality of reception antennas (also referredto as “reception array antenna”) to separate and receive multiplexedtransmission signals from reception signals. Through such processing,the MIMO radar performs array signal processing using these receptionsignals as a virtual reception array.

Further, in the MIMO radar, it is possible to enlarge the antennaaperture of the virtual reception array so as to enhance the angularresolution by appropriately arranging element spacings in transmissionand reception array antennas. Alternatively, the MIMO radar allowsreduction of sidelobes or grating lobes by more dense arrangement of theantenna spacings of the virtual reception array.

For example, Patent Literature (hereinafter, referred to as “PTL”) 1discloses a MIMO radar (hereinafter referred to as a “time-divisionmultiplexing MIMO radar”) that uses, as a multiplexing transmissionmethod for the MIMO radar, time-division multiplexing transmission bywhich signals are transmitted at transmission times shifted pertransmission antenna. The time-division multiplexing MIMO radar outputstransmission pulses, which are an example of transmission signals, whilesequentially switching the transmission antennas in a defined period.The time-division multiplexing MIMO radar receives, at a plurality ofreception antennas, signals that are the transmission pulses reflectedby an object, performs processing of correlating the reception signalswith the transmission pulses, and then performs, for example, spatialfast Fourier transform (FFT) processing (processing for estimation ofthe directions of arrival of the reflected waves).

The time-division multiplexing MIMO radar sequentially switches thetransmission antennas, from which the transmission signals (for example,the transmission pulses or radar transmission waves) are to betransmitted, in a defined period. The time-division multiplexing MIMOradar can thus extract propagation path responses indicated by theproduct (=Nt×Na) of number Nt of transmission antennas and number Na ofreception antennas, so as to perform the array signal processing usingthese Nt×Na reception signals as a virtual reception array. In otherwords, it is difficult to utilize the transmission antennas such thatthe number thereof is made greater than the number of transmissionantennas obtained by the transmission signals time-division multiplexedby switching of the transmission antennas (e.g., the number oftime-division multiplexing). For example, when the radar apparatustransmits a transmission signal using Nt transmission antennas by numberNt of time-division multiplexing, it is difficult to extract propagationpath responses that exceed (Nt×Na). Accordingly, when the number ofantennas is limited due to constraints such as the cost or installationlocation of the radar apparatus, the angular resolution or a sidelobereducing effect can be limited and it may be impossible to enhance theangular measurement performance.

Next, by way of example, attention will be paid to a method ofmultiplexing and transmitting transmission signals simultaneously from aplurality of transmission antennas.

Examples of the method for simultaneously multiplexing and transmittingtransmission signals from a plurality of transmission antennas include amethod (hereinafter referred to as Doppler multiplexing transmission)for transmitting signals such that a plurality of transmission signalscan be separated in the Doppler frequency domain on the receiver side(see, for example, NPL 2).

In the Doppler multiplexing transmission, transmission signalstransmitted from transmission antennas different from a referencetransmission antenna are, at a transmitter side, given respectiveDoppler shift amounts different from that given to a transmission signaltransmitted from the reference transmission antenna, and aresimultaneously transmitted from a plurality of transmission antennas(e.g., Nt transmission antennas). In the Doppler multiplexingtransmission, the signals received using a plurality of receptionantennas (e.g., Na reception antennas) are each filtered in the Dopplerfrequency domain, so that the transmission signals transmitted from thetransmission antennas are separated and received. Thus, the MIMO radarusing the Doppler multiplexing transmission (hereinafter, referred to as“Doppler multiplexing MIMO radar”) can extract propagation pathresponses indicated by the product (=Nt×Na) of number Nt of transmissionantennas and number Na of reception antennas, and performs array signalprocessing using these (Nt×Na) reception signals as a virtual receptionarray. In other words, it is difficult to utilize the transmissionantennas such that the number thereof is made greater than the number oftransmission antennas performing Doppler multiplexing transmission(e.g., the number of Doppler multiplexing). For example, when the radarapparatus transmits transmission signals using Nt transmission antennaswith number Nt of Doppler multiplexing, it is difficult to extractpropagation path responses that exceed (Nt×Na) in number.

Further, another method of multiplexing and transmitting transmissionsignals simultaneously from a plurality of transmission antennas is codemultiplexing transmission (see, for example, PTL 3). For example, a MIMOradar using the code multiplexing transmission (hereinafter, referred toas “code multiplexing MIMO radar”) performs code multiplexingtransmission from a plurality transmission antennas (e.g., Nttransmission antennas) by repeating, for each repeated transmission ofthe transmission signals (e.g., chirp signals), application of phasemodulation based on a code string (hereinafter, also referred to as acode or a code sequence) different for each transmission antenna.Further, the code multiplexing MIMO radar extracts range information ofcode-multiplexed reception signals by performing wave detectionprocessing on signals received using, for example, a plurality ofreception antennas (e.g., Na reception antennas). Further, the codemultiplexing MIMO radar performs, for example, on the range informationobtained for each repeated transmission of the transmission signals,Fourier transform processing in a velocity direction by dividing therange information into M pieces (for example, the code length of thecode string is used as M). The code multiplexing MIMO radar separatesthe code-multiplexed reception signals by applying phase correctionbased on detected velocity components to M results of the Fouriertransform processing in the velocity direction, and multiplying the Mresults by inverse code strings for separating code strings applied foreach transmission antenna.

Such a configuration of the code multiplexing MIMO radar allows the codemultiplexing MIMO radar to reduce mutual interference between thecode-multiplexed reception signals and separate the code-multiplexedreception signals, for example, even when the relative velocity betweena target and the code multiplexing MIMO radar is not zero. Thus, thecode multiplexing MIMO radar can extract propagation path responsesindicated by the product (=Nt×Na) of number Nt of transmission antennasand number Na of reception antennas, and performs array signalprocessing using these (Nt×Na) reception signals as a virtual receptionarray. In other words, it is difficult to utilize the transmissionantennas such that the number thereof is made greater than the number oftransmission antennas performing code multiplexing transmission (e.g.,the number of code multiplexing). For example, when the radar apparatustransmits transmission signals using Nt transmission antennas withnumber Nt of code multiplexing, it is difficult to extract propagationpath responses that exceed (Nt×Na) in number.

In view of the above, one exemplary embodiment according to the presentdisclosure will be described in relation to a method of utilizing thetransmission antennas such that the number thereof is made greater thanthe number of transmission antennas used for multiplexing transmission.In other words, the exemplary embodiment according to the presentdisclosure will be described in relation to, for example, a method ofextracting more than Nt×Na propagation path responses when a radarapparatus transmits a transmission signal using Nt transmission antennaswith number Nt of multiplexing. With this configuration, the radarapparatus of one exemplary embodiment according to the presentdisclosure can utilize more virtual reception antennas, and it is thuspossible to improve the angular measurement performance of the radarapparatus and improve the sensing accuracy for sensing a target object.

Embodiments of the present disclosure will be described in detail withreference to the drawings. In the embodiments, the same constituentelements are identified with the same numerals, and a descriptionthereof is omitted to avoid redundancy.

The following describes a configuration of a radar apparatus (in otherwords, MIMO radar configuration) having a transmitting branch in whichmultiplexed different transmission signals are simultaneously sent froma plurality of transmission antennas, and a receiving branch in whichthe transmission signals are separated and subjected to receptionprocessing.

Further, by way of example, a description will be given below of aconfiguration of a radar system using a frequency-modulated pulse wavesuch as a chirp pulse (e.g., also referred to as chirp pulsetransmission (fast chirp modulation)). However, the modulation scheme isnot limited to frequency modulation. For example, an exemplaryembodiment of the present disclosure is also applicable to a radarsystem that uses a pulse compression radar configured to transmit apulse train after performing phase modulation or amplitude modulation onthe pulse train.

Further, the radar apparatus performs Doppler multiplexing transmission,for example. In addition, in the Doppler multiplexing transmission, theradar apparatus multiplexes and transmits signals by encoding (forexample, performing code division multiplexing (CDM) on) the signals towhich different phase rotations (in other words, phase shifts), thenumber of which corresponds to the number of Doppler multiplexing, areapplied, (hereinafter, such signals are referred to as“Doppler-multiplexed transmission signals”) (hereinafter, suchmultiplexing is referred to as “Coded Doppler Multiplexing”).

[Configuration of Radar Apparatus]

Radar apparatus 10 in FIG. 1 includes radar transmitter (transmittingbranch) 100 and radar receiver (receiving branch) 200.

Radar transmitter 100 generates radar signals (radar transmissionsignals) and transmits the radar transmission signals in a definedtransmission period (hereinafter, referred to as “radar transmissionperiod”) using a transmission array antenna composed of a plurality oftransmission antennas 109 (for example, Nt transmission antennas).

Radar receiver 200 receives reflected wave signals, which are radartransmission signals reflected by a target object (target) (notillustrated), using a reception array antenna composed of a plurality ofreception antennas 202-1 to 202-Na. Radar receiver 200 performs signalprocessing on the reflected wave signals received at reception antennas202 to, for example, detect the presence or absence of the targetobject, or estimate the distances through which the reflected wavesignals arrive, the Doppler frequencies (in other words, the relativevelocities), and the directions of arrival, and outputs information onan estimation result (in other words, positioning information).

Note that, radar apparatus 10 may be mounted, for example, on a mobilebody such as a vehicle, and a positioning output of radar receiver 200(information on the estimation result) may, for example, be connected toan Electronic Control Unit (ECU) (not illustrated) such as an AdvancedDriver Assistance System (ADAS) or an autonomous driving system forenhancing the collision safety and utilized for a vehicle drive controlor alarm call control.

Radar apparatus 10 may also be mounted on a relatively high-altitudestructure (not illustrated), such as, for example, a roadside utilitypole or traffic lights. Radar apparatus 10 may also be utilized, forexample, as a sensor of a support system for enhancing the safety ofpassing vehicles or pedestrians, or as a sensor of a suspiciousintrusion prevention system (not illustrated). The positioning output ofradar receiver 200 may also be connected, for example, to a controldevice (not illustrated) in the support system or the suspiciousintrusion prevention system for enhancing safety and may be utilized foran alarm call control or an abnormality detection control. The use ofradar apparatus 10 is not limited to the above, and may also be used forother uses.

In addition, the target object is an object to be detected by radarapparatus 10.

Examples of the target object include vehicles (including four-wheel andtwo-wheel vehicles), a person, and a block or a curb.

[Configuration of Radar Transmitter 100]

Radar transmitter 100 includes radar transmission signal generator 101,phase rotation amount setter 105, phase rotators 108, and transmissionantennas 109.

Radar transmission signal generator 101 generates a radar transmissionsignal. Radar transmission signal generator 101 includes, for example,transmission signal generation controller 102, modulation signalgenerator 103, and Voltage Controlled Oscillator (VCO) 104. Theconstituent sections of radar transmission signal generator 101 will bedescribed below.

Transmission signal generation controller 102 sets, for example, atransmission signal generation timing for each radar transmissionperiod, and outputs information on the set transmission signalgeneration timing to modulation signal generator 103 and phase rotationamount setter 105 (e.g., Doppler shift setter 106). Here, the radartransmission period is represented by Tr.

Modulation signal generator 103 periodically generates, for example,saw-toothed modulation signals based on the information on thetransmission signal generation timing for each radar transmission periodTr inputted from transmission signal generation controller 102.

VCO 104 outputs, based on the modulation signals inputted frommodulation signal generator 103, frequency-modulated signals(hereinafter referred to as, for example, frequency chirp signals orchirp signals) to phase rotators 108 and radar receiver 200 (mixer 204described below) as the radar transmission signals (radar transmissionwaves) illustrated in FIG. 2.

Phase rotation amount setter 105 sets phase rotation amounts applied toradar signals for each radar transmission period Tr at phase rotators108 (in other words, phase rotation amounts corresponding to the codedDoppler multiplexing transmission) based on the information on thetransmission signal generation timing for each radar transmission periodTr inputted from transmission signal generation controller 102. Phaserotation amount setter 105 includes, for example, Doppler shift setter106 and encoder 107.

Doppler shift setter 106 sets phase rotation amounts that are applied tothe radar transmission signals (e.g., chirp signals) and that correspondto Doppler shift amounts, for example, based on the information on thetransmission signal generation timing for each radar transmission periodTr.

Encoder 107 sets a phase rotation amount corresponding to coding, forexample, based on the information on the transmission signal generationtiming for each radar transmission period Tr. Encoder 107 calculatesphase rotation amounts for phase rotators 108 based on, for example, thephase rotation amounts outputted from Doppler shift setter 106 and thephase rotation amount corresponding to coding, and outputs the phaserotation amounts to phase rotators 108. Further, encoder 107 outputs,for example, information on code sequences used for coding (for example,elements of orthogonal code sequences) to radar receiver 200 (forexample, output switch 209).

The number of coded Doppler multiplexing for Doppler multiplexed signalsthat is set by encoder 107 does not have to depend on the phase rotationamounts (Doppler shift amounts) of respective transmission antennas 109set by phase rotators 108. In other words, even when phase rotators 108sets the same phase rotation amount (Doppler shift amount) for a pair ofadjacent transmission antennas 109, encoder 107 may set the same numberof coded Doppler multiplexing or may set different values.

Phase rotators 108 apply the phase rotation amounts inputted fromencoder 107 to the chirp signals inputted from VCO 104 and outputs thesignals subjected to phase rotation to transmission antennas 109. Forexample, each of phase rotators 108 includes a phase shifter, a phasemodulator, and the like (not illustrated). The output signals of phaserotators 108 are amplified to a defined transmission power and areradiated respectively from transmission antennas 109 to space. In otherwords, radar transmission signals are multiplexed by application of thephase rotation amounts corresponding to the Doppler shift amounts andthe orthogonal code sequences and are transmitted from a plurality oftransmission antennas 109.

Next, an example method for phase rotation amount setter 105 to set thephase rotation amounts will be described.

Doppler shift setter 106 sets phase rotation amount φ_(ndm) for applyingDoppler shift amount DOP_(ndm) and outputs phase rotation amount φ_(ndm)to encoder 107. Here, ndm=1, . . . , N_(DM). N_(DM) denotes the setnumber of different Doppler shift amounts and is hereinafter referred toas the “number of Doppler multiplexing.”

In radar apparatus 10, since coding performed by encoder 107 is used forsome purposes, number N_(DM) of Doppler multiplexing may be set smallerthan number Nt of transmission antennas 109 used for multiplexingtransmission. Note that, number N_(DM) of Doppler multiplexing isgreater than or equal to 2.

Doppler shift amounts at equal intervals, or Doppler shift amounts atunequal intervals may, for example, be set as Doppler shift amountsDOP₁, DOP₂, . . . , and DOP_(N_DM) (“N_DM” is also represented as“N_(DM)”). Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM) maybe set to satisfy, for example, 0≤DOP₁, DOP₂, . . . ,DOP_(N_DM)<(1/TrL_(oc)) since the coding by encoder 107 described lateris used for some purposes. Alternatively, Doppler shift amounts DOP₁,DOP₂, . . . , and DOP_(N_DM), for example, may be set to satisfyExpression 1:

$\begin{matrix}{{\frac{- 1}{2T_{r}L_{oc}} \leq {DOP}_{1}},{DOP}_{2},\ldots\;,{{DOP}_{N\_{DM}} < {\frac{1}{2T_{r}L_{oc}}.}}} & \left( {{Expression}\mspace{14mu} 1} \right)\end{matrix}$

Further, for example, minimum Doppler shift interval Δf_(MinInterval)between Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM) maysatisfy following Expression 2. Note that, the Doppler shift intervalmay be defined as an absolute value of a difference between any two ofDoppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM). Here, Locrepresents the number of code elements. For example, Loc represents thecode length of a code used in encoder 107.

$\begin{matrix}{0 < {\Delta\; f_{MinInterval}} \leq \frac{1}{T_{r}N_{DM}L_{oc}}} & \left( {{Expression}\mspace{14mu} 2} \right)\end{matrix}$

Further, phase rotation amounts φ_(ndm) for applying Doppler shiftamounts DOP₁, DOP₂, . . . , and DOP_(N_DM) may, for example, be assignedas given by following Expression 3:

$\begin{matrix}{\phi_{ndm} = {2\pi\;{DOP}_{ndm}\text{/}{\left( \frac{1}{T_{r}L_{oc}} \right).}}} & \left( {{Expression}\mspace{14mu} 3} \right)\end{matrix}$

Note that, when the Doppler shift amounts at the equal interval ofΔf_(MinInterval) are set (hereinafter, such Doppler shift amounts arereferred to as “equal-interval Doppler shift amount setting”), phaserotation amounts φ_(ndm) for applying Doppler shift amounts DOP_(ndm)are assigned, for example, as given by following Expression 4:

$\begin{matrix}{\phi_{ndm} = {2{\pi\left( {{ndm} - 1} \right)}\Delta\; f_{MinInterval}\text{/}{\left( \frac{1}{T_{r}L_{oc}} \right).}}} & \left( {{Expression}\mspace{14mu} 4} \right)\end{matrix}$

Note that, as minimum Doppler shift interval Δf_(MinInterval) is madenarrower, the interference between Doppler multiplexed signals is morelikely to occur, and the target detection accuracy is more likely to bereduced (e.g., degraded). Thus, it is preferable that the intervalsbetween the Doppler shift amounts be widened as much as possible withinthe range satisfying the constraints of Expression 2. For example, whenthe equal sign holds true in Expression 2 (e.g.,Δf_(MinInterval)=1/(T_(r)N_(DM)L_(OC))), the intervals between theDoppler multiplexed signals in the Doppler domain can be maximized(hereinafter, referred to as “maximum equal-interval Doppler shiftamount setting”). In this case, a phase rotation range greater than orequal to 0 and less than 2π are equally divided into N_(DM) sub-ranges,and Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM) areassigned respective different phase rotation amounts. For example, phaserotation amount φ_(ndm) for applying Doppler shift amount DOP_(ndm) isassigned as given by following Expression 5. Note that, in thefollowing, the angle is expressed in radian.

$\begin{matrix}{\phi_{ndm} = \frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM}}} & \left( {{Expression}\mspace{14mu} 5} \right)\end{matrix}$

In Expression 5, for example, when number N_(DM) of Doppler multiplexingis 2, phase rotation amount φ₁ for applying Doppler shift amount DOP₁ is0, and phase rotation amount φ₂ for applying Doppler shift amount DOP₂is π. Likewise, in Expression 5, for example, when number N_(DM) ofDoppler multiplexing is 4, phase rotation amount φ₁ for applying Dopplershift amount DOP₁ is 0, phase rotation amount φ₂ for applying Dopplershift amount DOP₂ is π/2, phase rotation amount φ₃ for applying Dopplershift amount DOP₃ is π, and phase rotation amount φ₄ for applyingDoppler shift amount DOP₄ is 3π/2. In other words, intervals betweenphase rotation amounts φ_(ndm) for applying Doppler shift amountsDOP_(ndm) are equal intervals.

Note that, the assignment of the phase rotation amounts for applyingDoppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM) is not limitedto this assignment method. For example, the assignment of the phaserotation amounts given by Expression 5 may be shifted. For example, thephase rotation amounts may be assigned such that φ_(ndm)=2π(ndm)/N_(DM).Alternatively, phase rotation amounts φ₁, φ₂, . . . , and φ_(P_DM) maybe randomly assigned for Doppler shift amounts DOP₁, DOP₂, . . . , andDOP_(NDM) (where “N_DM” corresponds to N_(DM)) using an assignment tableof the phase rotation amounts.

In addition, in the equal-interval Doppler shift amount setting, whenthe denominator of phase rotation amount φ_(ndm) given by Expression 4is set to an integer and the phase rotation amounts are set to integervalues in units of Degree, it becomes easier to set the phase rotationamounts. For example, by settingΔf_(MinInterval)=1/T_(r)(N_(DM)+N_(int))L_(OC), the denominator of phaserotation amount φ_(ndm) given by Expression 4 is set to an integer valueas given by following Expression 6. Further, when N_(int) is set suchthat the value of the denominator (N_(DM)+N_(int)) in Expression 6 is adivisor of 360, the phase rotation amount is set to an integer value,and it becomes easier to set the phase rotation amounts.

$\begin{matrix}{\phi_{ndm} = \frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM} + N_{int}}} & \left( {{Expression}\mspace{14mu} 6} \right)\end{matrix}$

Here, N_(int) takes an integer value greater than or equal to 0. Forexample, when N_(int)=1 is set in the case of N_(DM)=7,ϕ_(ndm)=2π(ndm−1)/(N_(DM)+N_(int))=π(ndm−1)/4 holds. Accordingly, φ₁,φ₂, . . . , and φ_(N_DM) are integer values in units of Degree such as0°, 45°, 90°, 135°, . . . , and 270°, respectively, and it becomeseasier to set the phase rotation amounts.

Note that, when N_(int)=0 in Expression 6, the maximum equal-intervalDoppler shift amount setting is used.

Regarding phase rotation amounts φ₁, . . . , and φ_(N_DM) for applyingN_(DM) Doppler shift amounts inputted from Doppler shift setter 106,encoder 107 sets the phase rotation amounts based on one or a plurality(equal to or less than N_(CM)) of orthogonal code sequences. Further,encoder 107 sets the phase rotation amounts based on both the Dopplershift amounts and the orthogonal code sequences, for example, the “codedDoppler phase rotation amounts” for generating coded Doppler multiplexedsignals, and outputs the coded Doppler phase rotation amounts to phaserotators 108.

An example of the operation of encoder 107 will be described below.

For example, encoder 107 uses orthogonal code sequences with numberN_(CM) of codes (in other words, the number of code multiplexing) andwith code length Loc.

In the following, N_(CM) orthogonal code sequences with code length Locare denoted as Code_(ncm)={OC_(ncm)(1), OC_(ncm)(2), . . . ,OC_(ncm)(Loc)}. OC_(ncm)(noc) represents the noc-th code element inncm-th orthogonal code sequence Code_(ncm). Here, noc denotes the indexof a code element, and noc=1, . . . , Loc.

The orthogonal code sequences used in encoder 107 are, for example,codes that are orthogonal (uncorrelated) to one another. For example,the orthogonal code sequences may be Walsh-Hadamard codes. In this case,code length Loc used to generate orthogonal code sequences with numberN_(CM) of codes is given by following Expression 7.

[7]

Loc=2^(ceil[log) ² ^((N) ^(CM) ^()])  (Expression 7)

Here, ceil[x] is an operator (ceiling function) that outputs thesmallest integer greater than or equal to real number x.

For example, in a case where N_(CM)=2, code length Loc of Walsh-Hadamardcodes is 2, and the orthogonal code sequences are represented byCode₁={1, 1} and Code₂={1, −1}. Note that, when a code elementconstituting the orthogonal code sequences is 1, 1=exp(j0) holds trueand, thus, the phase thereof is 0. In addition, when a code elementconstituting the orthogonal code sequences is −1, −1=exp(jπ) holds trueand, thus, the phase thereof is π.

Further, for example, in a case where N_(CM)=4, code length Loc is 4,and the orthogonal code sequences are represented by Code₁={1, 1, 1, 1},Code₂={1, −1, 1, −1}, Code₃={1, 1, −1, −1}, and Code₄={1, −1, −1, 1}.

Note that, code elements constituting an orthogonal code sequence arenot limited to real numbers and may include complex number values. Forexample, orthogonal code sequence Code_(ncm) given by followingExpression 8 may be used. Here, ncm=1, . . . , N_(CM). In this case, thecode length used to generate orthogonal code sequences with numberN_(CM) of codes is represented by Loc=N_(CM).

$\begin{matrix}{{Code}_{ncm} = \left\{ {1,{\exp\left\lbrack {j\frac{2\pi}{N_{CM}}\left( {{ncm} - 1} \right)} \right\rbrack},{\exp\left\lbrack {j\frac{2\pi}{N_{CM}}2\left( {{ncm} - 1} \right)} \right\rbrack},\ldots\;,{\exp\left\lbrack {j\frac{2\pi}{N_{CM}}\left( {N_{CM} - 1} \right)\left( {{ncm} - 1} \right)} \right\rbrack}} \right\}} & \left( {{Expression}\mspace{14mu} 8} \right)\end{matrix}$

For example, in a case where N_(CM)=3, code length Loc is 3 (=N_(CM)),and encoder 107 generates orthogonal code sequences represented byCode₁={1, 1, 1}, Code₂={1, exp(j2π/3), exp(j4π/3)}, and Code₃={1,exp(−j2π/3), exp(−j4π/3)}.

Further, for example, in a case where N_(CM)=4, code length Loc is 4(=N_(CM)), and encoder 107 generates orthogonal code sequencesrepresented by Code₁={1, 1, 1, 1}, Code₂={1, j, −1, −j}, Code₃={1, −1,1, −1}, and Code₄={1, −j, −1, j}. Here, j is the imaginary unit.

In encoder 107, the number of code multiplexing (hereinafter referred toas the number of coded Doppler multiplexing) for encoding a Dopplermultiplexed signal using ndm-th Doppler shift amount DOP_(ndm) inputtedfrom Doppler shift setter 106 is represented by “N_(DOP_CODE)(ndm).”Here, ndm=1, . . . , N_(DM).

Encoder 107 sets number N_(DOP_CODE)(ndm) of coded Doppler multiplexingsuch that, for example, the sum of numbers N_(DOP_CODE)(1),N_(DOP_CODE)(2), . . . , and N_(DOP_CODE)(N_(DM)) of coded Dopplermultiplexing for encoding Doppler multiplexed signals is equal to numberNt of transmission antennas 109 used for multiplexing transmission. Inother words, encoder 107 sets number N_(DOP_CODE)(ndm) of coded Dopplermultiplexing so as to satisfy following Expression 9. This allows radarapparatus 10 to perform multiplexing transmission in the Doppler domainand in the code domain (hereinafter referred to as the coded Dopplermultiplexing transmission) using Nt transmission antennas 109.

$\begin{matrix}{{\sum\limits_{{ndm} = 1}^{N_{DM}}\;{N_{{DOP}\_{CODE}}({ndm})}} = {Nt}} & \left( {{Expression}\mspace{14mu} 9} \right)\end{matrix}$

Further, encoder 107 may set numbers N_(DOP_CODE)(1), N_(DOP_CODE)(2), .. . , and N_(DOP_CODE)(N_(DM)) of coded Doppler multiplexing, forexample, using the equal-interval Doppler shift amount setting includingthe maximum equal-interval Doppler shift amount setting, such thatdifferent numbers of coded Doppler multiplexing ranging from 1 throughN_(CM) are included. For example, encoder 107 does not use number N_(CM)of codes for all the numbers of coded Doppler multiplexing, but setsnumber N_(DOP_CODE)(ndm) of coded Doppler multiplexing corresponding toat least one Doppler shift amount DOP_(ndm) such that this number ofcoded Doppler multiplexing is smaller than N_(CM). Accordingly, among aplurality of combinations of the Doppler shift amounts DOP_(ndm) and theorthogonal code sequences, number N_(DOP_CODE)(ndm) of multiplexing(number of coded Doppler multiplexing) by the orthogonal code sequenceswhich is associated with at least one Doppler shift amount DOP_(ndm) maydiffer from the numbers of coded Doppler multiplexing associated withthe other Doppler shift amounts. In other words, encoder 107 sets thenumbers of coded Doppler multiplexing for the Doppler multiplexedsignals non-uniformly. With this setting, radar apparatus 10 canindividually separate and receive the coded Doppler multiplexed signalstransmitted from a plurality of transmission antennas 109 over a Dopplerrange of ±½Tr, for example, by aliasing judgement processing inreception processing described later.

Alternatively, encoder 107 may set numbers N_(DOP_CODE)(1),N_(DOP_CODE)(2), . . . , and N_(DOP_CODE)(N_(DM)) of coded Dopplermultiplexing, for example, using the equal-interval Doppler shift amountsetting of intervals narrower than the intervals of the maximumequal-interval Doppler shift amount setting, such that the same numberof coded Doppler multiplexing in the range of from 1 through N_(CM) isincluded. For example, encoder 107 may set number N_(CM) of codes forall the numbers of coded Doppler multiplexing. Accordingly, among aplurality of combinations of Doppler shift amounts DOP_(ndm) and theorthogonal code sequences, numbers N_(DOP_CODE)(ndm) of multiplexing(number of coded Doppler multiplexing) by the orthogonal code sequenceswhich are associated with Doppler shift amounts DOP_(ndm) may be thesame. In other words, encoder 107 sets the numbers of coded Dopplermultiplexing for the Doppler multiplexed signals uniformly. With thissetting, radar apparatus 10 can individually separate and receive thecoded Doppler multiplexed signals transmitted from a plurality oftransmission antennas 109 over a Doppler range of ±1/(2×Loc×Tr), forexample, by aliasing judgement processing in reception processingdescribed later.

Alternatively, encoder 107 may set numbers N_(DOP_CODE)(1),N_(DOP_CODE)(2), . . . , and N_(DOP_CODE)(N_(DM)) of coded Dopplermultiplexing, for example, using the maximum equal-interval Dopplershift amount setting such that the same number of coded Dopplermultiplexing in the range of from 1 through N_(CM) is included. Forexample, encoder 107 may set number N_(CM) of codes for all the numbersof coded Doppler multiplexing. In other words, encoder 107 sets thenumbers of coded Doppler multiplexing for the Doppler multiplexedsignals uniformly. In the case of this setting, for example, thealiasing judgement processing in the reception processing describedbelow is not applied. In addition, radar apparatus 10 can individuallyseparate and receive the coded Doppler multiplexed signals transmittedfrom a plurality of transmission antennas 109 over a Doppler range of±1/(2Loc×NEM×Tr), for example.

With respect to phase rotation amount φ_(ndm) for applying ndm-thDoppler shift amount DOP_(ndm), encoder 107 sets coded Doppler phaserotation amount ψ_(ndop_code(ndm), ndm)(m) for m-th transmission periodTr that is given by following Expression 10, and outputs coded Dopplerphase rotation amount ψ_(ndop_code(ndm), ndm)(m) to phase rotator 108:

$\begin{matrix}{{\psi_{{{{ndop}\_{code}}{({ndm})}},{ndm}}(m)} = {{{{floor}\left\lbrack \frac{\left( {m - 1} \right)}{Loc} \right\rbrack} \times \phi_{ndm}} + {{{angle}\left\lbrack {{OC}_{{{ndop}\_{code}}{({ndm})}}({OC\_ INDEX})} \right\rbrack}.}}} & \left( {{Expression}\mspace{14mu} 10} \right)\end{matrix}$

Here, the subscript “ndop_code(ndm)” represents an index less than orequal to number N_(DOP_CODE)(ndm) of coded Doppler multiplexing forphase rotation amount φ_(ndm) for applying Doppler shift amountDOP_(ndm). For example, ndop_code(ndm)=1, . . . , N_(DOP_CODE)(ndm).Here, angle[x] is an operator outputting the radian phase of real numberx, and for example, angle[1]=0, angle[−1]=π, angle[j]=π/2, andangle[j]=−π/2. In addition, floor[x] is an operator that outputs thelargest integer that does not exceed real number x. The character “j” isan imaginary unit.

For example, as given by Expression 10, coded Doppler phase rotationamount ψ_(ndop_code(ndm), ndm)(m) provides a constant phase rotationamount for applying Doppler shift amount DOP_(ndm) (for example, thefirst term in Expression 9) in the duration of Loc transmission periods(“Loc” is the code length used for coding), and applies a phase rotationamount corresponding to each of Loc code elementsOC_(ndop_code(ndm))(1), . . . , and OC_(ndop_code(ndm))(Loc) of codeCode_(ndop_code(ndm)) used for coding (the second term in Expression 9).

Further, encoder 107 outputs, in each transmission period (Tr),orthogonal code element index OC_INDEX to radar receiver 200 (outputswitch 209 described below). OC_INDEX represents an orthogonal codeelement index indicating an element of orthogonal code sequenceCode_(ndop_code(ndm)), and cyclically varies in the range of from 1 toLoc in each transmission period (Tr), as given by following Expression11:

[11]

OC_INDEX=mod(m−1,Loc)+1  (Expression 11).

Here, mod(x, y) denotes a modulo operator and is a function that outputsthe remainder after x is divided by y. Further, m=1, . . . , Nc. Ncdenotes the number of transmission periods used for radar positioning(hereinafter referred to as “radar-transmission-signal transmissiontimes”). In addition, radar-transmission-signal transmission times Nc isset to an integer multiple of Loc (by a factor of Ncode). For example,Nc=Loc×Ncode.

Next, an example method by encoder 107 for setting numbersN_(DOP_CODE)(ndm) of coded Doppler multiplexing for Doppler multiplexedsignals non-uniformly will be described.

For example, encoder 107 sets number N_(CM) of orthogonal code sequences(in other words, the number of code multiplexing or the number of codes)satisfying the condition below. For example, number N_(CM) of orthogonalcode sequences and number N_(DM) of Doppler multiplexing satisfy thefollowing relationship for number Nt of transmission antennas 109 usedfor multiplexing transmission:

(Number N _(CM) of orthogonal code sequences)×(Number N _(DM) of Dopplermultiplexing)>Number Nt of transmission antennas used for multiplexingtransmission.

For example, among numbers N_(CM) of orthogonal code sequences andnumbers N_(DM) of Doppler multiplexing satisfying the above-describedcondition, the use of a combination yielding a smaller product(N_(CM)×N_(DM)) is desirable in terms of both characteristics andcomplexity of circuit configuration. Note that among numbers N_(CM) oforthogonal code sequences and numbers N_(DM) of Doppler multiplexingsatisfying the above-described condition, a combination having a smallervalue of the product (N_(CM)×N_(DM)) is not limitative, and any othercombination may be applied.

For example, in a case where Nt=3, the combination of N_(DM)=2 andN_(CM)=2 is desirable.

In this case, the assignment of Doppler shift amounts DOP₁ and DOP₂ andorthogonal codes Code₁ and Code₂ is determined in accordance with thesetting of N_(DOP_CODE)(1) and N_(DOP_CODE)(2), for example, asillustrated at (a) and (b) in FIG. 3. In FIG. 3, white circles (“∘”)represent Doppler shift amounts and orthogonal codes used, and crosses(“x”) represent Doppler shift amounts and orthogonal codes not used (thesame applies to the following description).

For example, (a) in FIG. 3 illustrates an example of N_(DOP_CODE)(1)=2and N_(DOP_CODE)(2)=1, and (b) in FIG. 3 illustrates an example ofN_(DOP_CODE)(1)=1 and N_(DOP_CODE)(2)=2.

Note that, in FIG. 3, Code₁ is used for the Doppler shift amount (e.g.,DOP₂ at (a) in FIG. 3 and DOP₁ at (b) in FIG. 3) corresponding to numberN_(DOP_CODE)(ndm)=1 of coded Doppler multiplexing, but the presentdisclosure is not limited thereto. For example, in the case ofN_(DOP_CODE)(1)<N_(CM), or N_(DOP_CODE)(2)<N_(CM), Code₂ instead ofCode₁ may be used for the Doppler shift amount (e.g., DOP₂ at (a) inFIG. 4 and DOP₁ at (b) in FIG. 4) corresponding to N_(DOP_CODE)(ndm)=1as illustrated in FIG. 4.

Further, for example, in a case where Nt=4 or 5, the combination ofN_(DM)=3 and N_(CM)=2 or the combination of N_(DM)=2 and N_(CM)=3 isdesirable.

By way of example, FIG. 5 illustrate a case where Nt=4, N_(DM)=3, andN_(CM)=2. For example, the assignment of Doppler shift amounts DOP₁,DOP₂, and DOP₃ and orthogonal codes Code₁ and Code₂ is determined inaccordance with the setting of N_(DOP_CODE)(1), N_(DOP_CODE)(2), andN_(DOP_CODE)(3) as illustrated in FIG. 5.

For example, (a) in FIG. 5 illustrates an example whereN_(DOP_CODE)(1)=2, N_(DOP_CODE)(2)=1, and N_(DOP_CODE)(3)=1, (b) in FIG.5 illustrates an example where N_(DOP_CODE)(1)=1, N_(DOP_CODE)(2)=2, andN_(DOP_CODE)(3)=1, and (c) in FIG. 5 illustrates an example whereN_(DOP_CODE)(1)=1, N_(DOP_CODE)(2)=1, and N_(DOP_CODE)(3)=2.

Note that, in FIG. 5, Code₁ is used for the Doppler shift amountscorresponding to number N_(DOP_CODE)(ndm)=1 of coded Dopplermultiplexing, but the present disclosure is not limited thereto. Forexample, for settings in which the numbers of coded Doppler multiplexingare each smaller than N_(CM), Code₂ may be used in place of Code₁ asillustrated at (a) in FIG. 6, or both Code₁ and Code₂ may be used asillustrated at (b) or (c) in FIG. 6.

By way of another example, FIG. 7 illustrates a case where Nt=4,N_(DM)=2, and N_(CM)=3. For example, the assignment of Doppler shiftamounts DOP₁ and DOP₂ and orthogonal codes Code₁, Code₂, and Code₃ isdetermined in accordance with the setting of N_(DOP_CODE)(1) andN_(DOP_CODE)(2) as illustrated in FIG. 7.

For example, (a) in FIG. 7 illustrates an example whereN_(DOP_CODE)(1)=3 and N_(DOP_CODE)(2)=1, and (b) in FIG. 7 illustratesan example where N_(DOP_CODE)(1)=1 and N_(DOP_CODE)(2)=3.

Note that, in FIG. 7, Code₁ is used for the Doppler shift amountscorresponding to number N_(DOP_CODE)(ndm)=1 of coded Dopplermultiplexing, but the present disclosure is not limited thereto. Forexample, when N_(DOP_CODE)(1)<N_(CM) or N_(DOP_CODE)(2)<N_(CM), Code₂may be used in place of Code₁ as illustrated at (a) in FIG. 8, and Code₃may be used in place of Code₁ as illustrated at (b) in FIG. 8.

As in the case of N_(DOP_CODE)(1)=3 and N_(DOP_CODE)(2)=1, or the caseof N_(DOP_CODE)(1)=1 and N_(DOP_CODE)(2)=3 as illustrated in FIG. 7,numbers N_(DOP_CODE) of coded Doppler multiplexing are set non-uniformlyfor Doppler shift amounts DOP₁ and DOP₂. In such settings, the Dopplerfrequency range can be equivalent to, for example, the maximum Dopplervelocity at the time of single-antenna transmission (the details will bedescribed below).

Further, for example, in a case where Nt=6 or 7, the combination ofN_(DM)=4 and N_(CM)=2 or the combination of N_(DM)=2 and N_(CM)=4 isdesirable.

By way of example, FIG. 9 illustrates a case where Nt=6, N_(DM)=4, andN_(CM)=2. For example, the assignment of Doppler shift amounts DOP₁,DOP₂, DOP₃, and DOP₄ and orthogonal codes Code₁ and Code₂ is determinedin accordance with the setting of N_(DOP_CODE)(1), N_(DOP_CODE)(2),N_(DOP_CODE)(3), and N_(DOP_CODE)(4) as illustrated in FIG. 9.

For example, (a) in FIG. 9 illustrates an example whereN_(DOP_CODE)(1)=N_(DOP_CODE)(2)=2 and N_(DOP_CODE)(3)=N_(DOP_CODE)(4)=1,and (b) in FIG. 9 illustrates an example whereN_(DOP_CODE)(1)=N_(DOP_CODE)(3)=2 and N_(DOP_CODE)(2)=N_(DOP_CODE)(4)=1.

Note that, in FIG. 9, Code₁ is used for the Doppler shift amountscorresponding to number N_(DOP_CODE)(ndm)=1 of coded Dopplermultiplexing, but the present disclosure is not limited thereto. Forexample, for settings in which the numbers of coded Doppler multiplexingare each smaller than N_(CM), Code₂ may be used in place of Code₁ asillustrated in at (a) in FIG. 10, or both Code₁ and Code₂ may be used asillustrated at (b) in FIG. 10.

Further, for example, as illustrated in FIG. 9, in a case where Nt=6,N_(DM)=4, and N_(CM)=2, there are two Doppler shift amounts that do notuse all the codes. Further, for example, in the case of N_(DM)=4, inrespect of the combinations of Doppler shift amounts that do not use allthe codes, there are six combinations (=₄C₂) of two Doppler shiftamounts selected from four Doppler shift amounts, and in each of the sixcombinations, there are four combinations (=N_(CM)×N_(CM)) of codesused. Accordingly, in a case where Nt=6, N_(DM)=4, and N_(CM)=2, thereis a total of 24 combinations of Doppler shift amounts DOP andorthogonal codes Code assigned.

Likewise, for example, in a case where Nt=8, the combination of N_(DM)=3and N_(CM)=3 or the combination of N_(DM)=5 and N_(CM)=2 is desirable.For example, in a case where Nt=9, the combination of N_(DM)=5 andN_(CM)=2 is desirable. For example, in a case where Nt=10, thecombination of N_(DM)=6 and N_(CM)=2 or the combination of N_(DM)=4 andN_(CM)=3 is desirable. For example, in a case where Nt=12, thecombination of N_(DM)=5 and N_(CM)=3 or the combination of N_(DM)=4 andN_(CM)=4 is desirable. Note that, number Nt of transmission antennas 109is not limited to that in the examples described above, and an exemplaryembodiment of the present disclosure is also applicable in the casewhere Nt=11 or more.

Next, an example method by encoder 107 for setting numbersN_(DOP_CODE)(ndm) of coded Doppler multiplexing uniformly for Dopplermultiplexed signals will be described.

Note that, for the method by encoder 107 for setting numbersN_(DOP_CODE)(ndm) of coded Doppler multiplexing uniformly for Dopplermultiplexed signals, the use of a combination having a smaller product(N_(CM)×N_(DM)) from among numbers N_(CM) of orthogonal code sequencesand numbers N_(DM) of Doppler multiplexing satisfying the followingcondition is desirable in terms of both characteristics and complexityof circuit configuration. However, the present disclosure is not limitedto the combination having a smaller value of the product(N_(CM)×N_(DM)), but other combinations may also be applicable.

For example, encoder 107 sets number N_(CM) of orthogonal code sequences(in other words, the number of code multiplexing or the number of codes)satisfying the condition below. For example, number N_(CM) of orthogonalcode sequences and number N_(DM) of Doppler multiplexing satisfy thefollowing relationship for number Nt of transmission antennas 109 usedfor multiplexing transmission:

(Number N _(CM) of orthogonal code sequences)×(Number N _(DM) of Dopplermultiplexing)=Number Nt of transmission antennas used for multiplexingtransmission.

For example, in a case where Nt=4, the combination of N_(DM)=2 andN_(CM)=2 is desirable. For example, in a case where Nt=6, thecombination of N_(DM)=2 and N_(CM)=3 or the combination of N_(DM)=3 andN_(CM)=2 is desirable. For example, in a case where Nt=8, thecombination of N_(DM)=4 and N_(CM)=2 or the combination of N_(DM)=2 andN_(CM)=4 is desirable. For example, in a case where Nt=9, thecombination of N_(DM)=3 and N_(CM)=3 is desirable. For example, in acase where Nt=10, the combination of N_(DM)=2 and N_(CM)=5 or thecombination of N_(DM)=5 and N_(CM)=2 is desirable. Further, for example,in a case where Nt=12, the combination of N_(DM)=2 and N_(CM)=6, thecombination of N_(DM)=6 and N_(CM)=2, the combination of N_(DM)=3 andN_(CM)=4, or the combination of N_(DM)=4 and N_(CM)=3 is desirable.

Note that, number Nt of transmission antennas 109 is not limited tothose in the examples described above, and the exemplary embodiment ofthe present disclosure is applicable to other numbers. In this case, inorder to satisfy the combination of integers satisfying number N_(CM) oforthogonal code sequences>1 and number N_(DM) of Doppler multiplexing>1,and to satisfy (number N_(CM) of orthogonal code sequences)×(numberN_(DM) of Doppler multiplexing)=number Nt of transmission antennas usedfor multiplexing transmission, number Nt of transmission antennas usedfor multiplexing transmission may be set to 4 or more, and to satisfythe above condition.

Further, the example method by encoder 107 for setting numbersN_(DOP_CODE)(ndm) of coded Doppler multiplexing uniformly for Dopplermultiplexed signals is not limited to the above, and a larger N_(CM) maybe set in the above-described combinations.

For example, in a case where Nt=4, a combination satisfying N_(DM)=2 andN_(CM)≥2 is possible.

Further, for example, in the case of Nt=6, a combination of N_(DM)=2 andN_(CM)≥3, or a combination of N_(DM)=3 and N_(CM)≥2 is possible.

Further, for example, in the case of Nt=8, a combination of N_(DM)=4 andN_(CM)≥2, or a combination of N_(DM)=2 and N_(CM)≥4 is possible.

Further, for example, in the case of Nt=9, a combination of N_(DM)=3 andN_(CM)≥3 is possible.

Further, for example, in the case of Nt=10, a combination of N_(DM)=2and N_(CM)≥5, or a combination of N_(DM)=5 and N_(CM)≥2 is possible.

Further, for example, in the case of Nt=12, a combination of N_(DM)=2and N_(CM)≥6, a combination of N_(DM)=6 and N_(CM)≥2, a combination ofN_(DM)=3 and N_(CM)≥4, or a combination of N_(DM)=4 and N_(CM)≥3 ispossible.

For example, in a case where Nt=4, N_(DM)=2, and N_(CM)=3, ifN_(DOP_CODE)(1)=2 and N_(DOP_CODE)(2)=2 are set as illustrated in FIG.11, numbers N_(DOP_CODE) of coded Doppler multiplexing are uniformly setfor Doppler shift amounts DOP₁ and DOP₂. In this setting, for example,it is assumed that the same codes (for example, Code₁ and Code₂) areassigned for Doppler shift amounts DOP₁ and DOP₂ as illustrated at (a)in FIG. 11, or different codes are assigned for Doppler shift amountsDOP₁ and DOP₂ as illustrated at (b) in FIG. 11.

Next, an example of setting of coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm)(m) will be described.

For example, a description will be given of a case where in encoder 107,number Nt of transmission antennas used for multiplexing transmission is3, number N_(DM) of Doppler multiplexing is 2, and number N_(CM) of codemultiplexing is 2, and orthogonal code sequences Code₁={1, 1} andCode₂={1, −1} with code length Loc=2 are used. In this case, forexample, when numbers N_(DOP_CODE)(1) and N_(DOP_CODE)(2) of codedDoppler multiplexing are 1 and 2, encoder 107 sets coded Doppler phaserotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) given byfollowing Expressions 12 to 14 and outputs coded Doppler phase rotationamounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) to phase rotators 108:

$\begin{matrix}{{\left\{ {{\psi_{1,1}(1)},{\psi_{1,1}(2)},{\psi_{1,1}(3)},{\psi_{1,1}(4)},{\psi_{1,1}(5)},{\psi_{1,1}(6)},{\psi_{1,1}(7)},{\psi_{1,1}(8)},\ldots}\; \right\} = \left\{ {0,0,\phi_{1},\phi_{1},{2\phi_{1}},{2\phi_{1}},{3\phi_{1}},{3\phi_{1}},\ldots}\; \right\}};} & \left( {{Expression}\mspace{14mu} 12} \right) \\{{\left\{ {{\psi_{1,2}(1)},{\psi_{1,2}(2)},{\psi_{1,2}(3)},{\psi_{1,2}(4)},{\psi_{1,2}(5)},{\psi_{1,2}(6)},{\psi_{1,2}(7)},{\psi_{1,2}(8)},\ldots}\; \right\} = \left\{ {0,0,\phi_{1},\phi_{1},{2\phi_{1}},{2\phi_{1}},{3\phi_{1}},{3\phi_{1}},\ldots}\; \right\}};} & \left( {{Expression}\mspace{14mu} 13} \right) \\{\left\{ {{\psi_{2,2}(1)},{\psi_{2,2}(2)},{\psi_{2,2}(3)},{\psi_{2,2}(4)},{\psi_{2,2}(5)},{\psi_{2,2}(6)},{\psi_{2,2}(7)},{\psi_{2,2}(8)},\ldots}\; \right\} = {\left\{ {0,\pi,\phi_{2},{\phi_{2} + \pi},{2\phi_{2}},{{2\phi_{2}} + \pi},{3\phi_{2}},{{3\phi_{2}} + \pi},\ldots}\; \right\}.}} & \left( {{Expression}\mspace{14mu} 14} \right)\end{matrix}$

Here, as an example, φ_(ndm)=2π(ndm−1)/N_(DM) in Expression 5 is used asthe phase rotation amount for applying Doppler shift amount DOP_(ndm),and phase rotation amount φ₁=0 for applying Doppler shift amount DOP₁and phase rotation amount φ₂=π for applying Doppler shift amount DOP₂are used. In this case, encoder 107 sets coded Doppler phase rotationamounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) given by followingExpressions 15 to 17 and outputs coded Doppler phase rotation amountsψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) to phase rotators 108. Here,m=1, . . . , Nc. Here, a modulo operation for 2π is performed, andresults are expressed in radians ranging from 0 to less than 2π (thesame applies to the following description).

$\begin{matrix}{\left\{ {{\psi_{1,1}(1)},{\psi_{1,1}(2)},{\psi_{1,1}(3)},{\psi_{1,1}(4)},{\psi_{1,1}(5)},{\psi_{1,1}(6)},{\psi_{1,1}(7)},{\psi_{1,1}(8)},\ldots}\; \right\} = \left\{ {0,0,0,0,0,0,0,0,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 15} \right) \\{\left\{ {{\psi_{1,2}(1)},{\psi_{1,2}(2)},{\psi_{1,2}(3)},{\psi_{1,2}(4)},{\psi_{1,2}(5)},{\psi_{1,2}(6)},{\psi_{1,2}(7)},{\psi_{1,2}(8)},\ldots}\; \right\} = \left\{ {0,0,\pi,\pi,0,0,\pi,\pi,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 16} \right) \\{\left\{ {{\psi_{2,2}(1)},{\psi_{2,2}(2)},{\psi_{2,2}(3)},{\psi_{2,2}(4)},{\psi_{2,2}(5)},{\psi_{2,2}(6)},{\psi_{2,2}(7)},{\psi_{2,2}(8)},\ldots}\; \right\} = \left\{ {0,\pi,\pi,0,0,\pi,\pi,0,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 17} \right)\end{matrix}$

As given by Expressions 15 to 17, when the phase rotation amounts areset to φ_(ndm)=2π(ndm−1)/N_(DM), into which 2π is equally divided, codedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m)are changed in transmission periods given by N_(DM)×N_(CM)=2×2=4.

As another example, φ_(ndm)=2π(ndm)/N_(DM) may be used as the phaserotation amount for applying Doppler shift amount DOP_(ndm), and phaserotation amount φ₁=π for applying Doppler shift amount DOP₁ and phaserotation amount φ₂=0 for applying Doppler shift amount DOP₂ may be set.In this case, encoder 107 sets coded Doppler phase rotation amountsψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) as given by followingExpressions 18 to 20 and outputs coded Doppler phase rotation amountsψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) to phase rotators 108. Here,m=1, . . . , Nc.

$\begin{matrix}{\left\{ {{\psi_{1,1}(1)},{\psi_{1,1}(2)},{\psi_{1,1}(3)},{\psi_{1,1}(4)},{\psi_{1,1}(5)},{\psi_{1,1}(6)},{\psi_{1,1}(7)},{\psi_{1,1}(8)},\ldots}\; \right\} = \left\{ {0,0,\pi,\pi,0,0,\pi,\pi,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 18} \right) \\{\left\{ {{\psi_{1,2}(1)},{\psi_{1,2}(2)},{\psi_{1,2}(3)},{\psi_{1,2}(4)},{\psi_{1,2}(5)},{\psi_{1,2}(6)},{\psi_{1,2}(7)},{\psi_{1,2}(8)},\ldots}\; \right\} = \left\{ {0,0,0,0,0,0,0,0,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 19} \right) \\{\left\{ {{\psi_{2,2}(1)},{\psi_{2,2}(2)},{\psi_{2,2}(3)},{\psi_{2,2}(4)},{\psi_{2,2}(5)},{\psi_{2,2}(6)},{\psi_{2,2}(7)},{\psi_{2,2}(8)},\ldots}\; \right\} = \left\{ {0,\pi,0,\pi,0,\pi,0,\pi,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 20} \right)\end{matrix}$

As given by Expressions 15 to 17 or Expressions 18 to 20, the number ofphases (for example, two phases of 0 and π) used for the phase rotationamounts (for example, the phase rotation amounts for applying theDoppler shift amounts) is smaller than number Nt=3 of transmissionantennas 109 used for multiplexing transmission. In other words, asgiven by Expressions 15 to 17 or Expressions 18 to 20, the number ofphases (for example, two phases of 0 and π) used for the phase rotationamounts for applying the Doppler shift amounts is equal to numberN_(DM)=2 of Doppler shift amounts used for multiplexing transmission (inother words, the number of Doppler multiplexing).

In addition, for example, a description will be given of a case where inencoder 107, number Nt of transmission antennas used for multiplexingtransmission is 6, number N_(DM) of Doppler multiplexing is 4, andnumber N_(CM) of code multiplexing is 2, and orthogonal code sequencesCode₁={1, 1} and Code₂={1, −1} with code length Loc=2 are used. In thiscase, for example, if numbers N_(DOP_CODE)(1), N_(DOP_CODE)(2),N_(DOP_CODE)(3), and N_(DOP_CODE)(4) of coded Doppler multiplexing are1, 1, 2, and 2, respectively, encoder 107 sets coded Doppler phaserotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ1, 3(m), ψ2, 3(m), ψ1, 4(m),and ψ2, 4(m) given by following Expressions 21 to 26 and outputs codedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ1, 3(m), ψ2,3(m), ψ1, 4(m), and ψ2, 4(m) to phase rotators 108. Here, m=1, . . . ,Nc.

$\begin{matrix}{\left\{ {{\psi_{1,1}(1)},{\psi_{1,1}(2)},{\psi_{1,1}(3)},{\psi_{1,1}(4)},{\psi_{1,1}(5)},{\psi_{1,1}(6)},{\psi_{1,1}(7)},{\psi_{1,1}(8)},\ldots}\; \right\} = \left\{ {0,0,\phi_{1},\phi_{1},{2\phi_{1}},{2\phi_{1}},{3\phi_{1}},{3\phi_{1}},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 21} \right) \\{\left\{ {{\psi_{1,2}(1)},{\psi_{1,2}(2)},{\psi_{1,2}(3)},{\psi_{1,2}(4)},{\psi_{1,2}(5)},{\psi_{1,2}(6)},{\psi_{1,2}(7)},{\psi_{1,2}(8)},\ldots}\; \right\} = \left\{ {0,0,\phi_{2},\phi_{2},{2\phi_{2}},{2\phi_{2}},{3\phi_{2}},{3\phi_{2}},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 22} \right) \\{\left\{ {{\psi_{1,3}(1)},{\psi_{1,3}(2)},{\psi_{1,3}(3)},{\psi_{1,3}(4)},{\psi_{1,3}(5)},{\psi_{1,3}(6)},{\psi_{1,3}(7)},{\psi_{1,3}(8)},\ldots}\; \right\} = \left\{ {0,0,\phi_{3},\phi_{3},{2\phi_{3}},{2\phi_{3}},{3\phi_{3}},{3\phi_{3}},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 23} \right) \\{\left\{ {{\psi_{2,3}(1)},{\psi_{2,3}(2)},{\psi_{2,3}(3)},{\psi_{2,3}(4)},{\psi_{2,3}(5)},{\psi_{2,3}(6)},{\psi_{2,3}(7)},{\psi_{2,3}(8)},\ldots}\; \right\} = \left\{ {0,0,\phi_{3},\phi_{3},{2\phi_{3}},{2\phi_{3}},{3\phi_{3}},{3\phi_{3}},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 24} \right) \\{\left\{ {{\psi_{1,4}(1)},{\psi_{1,4}(2)},{\psi_{1,4}(3)},{\psi_{1,4}(4)},{\psi_{1,4}(5)},{\psi_{1,4}(6)},{\psi_{1,4}(7)},{\psi_{1,4}(8)},\ldots}\; \right\} = \left\{ {0,0,\phi_{4},\phi_{4},{2\phi_{4}},{2\phi_{4}},{3\phi_{4}},{3\phi_{4}},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 25} \right) \\{\left\{ {{\psi_{2,4}(1)},{\psi_{2,4}(2)},{\psi_{2,4}(3)},{\psi_{2,4}(4)},{\psi_{2,4}(5)},{\psi_{2,4}(6)},{\psi_{2,4}(7)},{\psi_{2,4}(8)},\ldots}\; \right\} = \left\{ {0,\pi,\phi_{4},{\phi_{4} + \pi},{2\phi_{4}},{{2\phi_{4}} + \pi},{3\phi_{4}},{{3\phi_{4}} + \pi},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 26} \right)\end{matrix}$

Here, as an example, φ_(ndm)=2π(ndm−1)/N_(DM) is used as the phaserotation amount for applying Doppler shift amount DOP_(ndm), and phaserotation amount φ₁=0 for applying Doppler shift amount DOP₁, phaserotation amount φ₂=π/2 for applying Doppler shift amount DOP₂, phaserotation amount φ₃=π for applying Doppler shift amount DOP₃, and phaserotation amount φ₄=3π/2 for applying Doppler shift amount DOP₄ are used.In this case, encoder 107 sets coded Doppler phase rotation amountsψ_(1, 1)(m), ψ_(1, 2)(m), ψ1, 3(m), ψ2, 3(m), ψ1, 4(m), and ψ2, 4(m)given by following Expressions 27 to 32 and outputs coded Doppler phaserotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ1, 3(m), ψ2, 3(m), ψ1, 4(m),and ψ2, 4(m) to phase rotators 108. Here, m=1, . . . , Nc.

$\begin{matrix}{\left\{ {{\psi_{1,1}(1)},{\psi_{1,1}(2)},{\psi_{1,1}(3)},{\psi_{1,1}(4)},{\psi_{1,1}(5)},{\psi_{1,1}(6)},{\psi_{1,1}(7)},{\psi_{1,1}(8)},\ldots}\; \right\} = \left\{ {0,0,0,0,0,0,0,0,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 27} \right) \\{\left\{ {{\psi_{1,2}(1)},{\psi_{1,2}(2)},{\psi_{1,2}(3)},{\psi_{1,2}(4)},{\psi_{1,2}(5)},{\psi_{1,2}(6)},{\psi_{1,2}(7)},{\psi_{1,2}(8)},\ldots}\; \right\} = \left\{ {0,0,\frac{\pi}{2},\frac{\pi}{2},\pi,\pi,\frac{3\pi}{2},\frac{3\pi}{2},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 28} \right) \\{\left\{ {{\psi_{1,3}(1)},{\psi_{1,3}(2)},{\psi_{1,3}(3)},{\psi_{1,3}(4)},{\psi_{1,3}(5)},{\psi_{1,3}(6)},{\psi_{1,3}(7)},{\psi_{1,3}(8)},\ldots}\; \right\} = \left\{ {0,0,\pi,\pi,0,0,\pi,\pi,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 29} \right) \\{\left\{ {{\psi_{2,3}(1)},{\psi_{2,3}(2)},{\psi_{2,3}(3)},{\psi_{2,3}(4)},{\psi_{2,3}(5)},{\psi_{2,3}(6)},{\psi_{2,3}(7)},{\psi_{2,3}(8)},\ldots}\; \right\} = \left\{ {0,\pi,\pi,0,0,\pi,\pi,0,\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 30} \right) \\{\left\{ {{\psi_{1,4}(1)},{\psi_{1,4}(2)},{\psi_{1,4}(3)},{\psi_{1,4}(4)},{\psi_{1,4}(5)},{\psi_{1,4}(6)},{\psi_{1,4}(7)},{\psi_{1,4}(8)},\ldots}\; \right\} = \left\{ {0,0,\frac{3\pi}{2},\frac{3\pi}{2},\pi,\pi,\frac{\pi}{2},\frac{\pi}{2},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 31} \right) \\{\left\{ {{\psi_{2,4}(1)},{\psi_{2,4}(2)},{\psi_{2,4}(3)},{\psi_{2,4}(4)},{\psi_{2,4}(5)},{\psi_{2,4}(6)},{\psi_{2,4}(7)},{\psi_{2,4}(8)},\ldots}\; \right\} = \left\{ {0,\pi,\frac{3\pi}{2},\frac{\pi}{2},\pi,0,\frac{\pi}{2},\frac{3\pi}{2},\ldots}\; \right\}} & \left( {{Expression}\mspace{14mu} 32} \right)\end{matrix}$

As given by Expressions 27 to 32, when the phase rotation amounts areset to φ_(ndm)=2π(ndm−1)/N_(DM), into which a is equally divided, codedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ1, 3(m), ψ2,3(m), ψ1, 4(m), and ψ2, 4(m) are changed in transmission periods givenby N_(DM)×N_(CM)=4×2=8.

Further, as illustrated in Expressions 27 to 32, the number of phasesused for the phase rotation amounts (e.g., the phase rotation amountsfor applying the Doppler shift amounts) (e.g., four phases of 0, π/2, π,and 3π/2) is less than number Nt=6 of transmission antenna 109 used formultiplexing transmission. In other words, as given by Expressions 27 to32, the number of phases (for example, four phases of 0, π/2, π, and3π/2) used for the phase rotation amounts for applying the Doppler shiftamounts is equal to number N_(DM)=4 of Doppler shift amounts used formultiplexing transmission (in other words, the number of Dopplermultiplexing).

The description has been given of, as examples, the settings of phaserotation amounts in a case where number Nt of transmission antennas 109is 3 and number N_(DM) of Doppler multiplexing is 2 and in a case wherenumber Nt of transmission antennas 109 is 6 and number N_(DM) of Dopplermultiplexing is 4. However, number Nt of transmission antennas 109 andnumber N_(DM) of Doppler multiplexing are not limited to the valuesdescribed above. For example, the number of phases used for the phaserotation amounts may be set smaller than number Nt of transmissionantennas 109 used for multiplexing transmission, regardless of number Ntof transmission antennas 109. Further, the number of phases used for thephase rotation amounts for applying the Doppler shift amounts may beequal to number N_(DM) of Doppler shift amounts used for multiplexingtransmission.

Further, the above example has been described with respect to themaximum equal-interval Doppler shift amount setting of the phaserotation amounts. However, the setting of the phase rotation amounts isnot limited thereto, and the equal-interval Doppler shift amount settingof the phase rotation amounts (for example, Expression 6) may be used.

The foregoing description has been given of the method for phaserotation amount setter 105 to set the phase rotation amounts.

In FIG. 1, phase rotators 108 apply the phase rotation amounts in eachtransmission period Tr to the chirp signals inputted from radartransmission signal generator 101, based on coded Doppler phase rotationamounts ψ_(ndop_code(ndm), ndm)(m) set by phase rotation amount setter105. Here, ndm=1, . . . , N_(DM), and ndop_code(ndm)=1, . . . ,N_(DOP_CODE)(ndm).

The sum of numbers N_(DOP_CODE)(1), N_(DOP_CODE)(2), . . . , andN_(DOP_CODE)(N_(DM)) of coded Doppler multiplexing is set to be equal tonumber Nt of transmission antennas 109, and Nt coded Doppler phaserotation amounts are respectively inputted to Nt phase rotators 108.

Each of Nt phase rotators 108 applies, in each transmission period Tr,inputted coded Doppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m)to a chirp signal inputted from radar transmission signal generator 101.The outputs of Nt phase rotators 108 (referred to as, for example, codedDoppler multiplexed signals) are amplified to a defined transmissionpower and are then radiated into space from Nt transmission antennas 109of a transmission array antenna section.

In the following, phase rotator 108 that applies coded Doppler phaserotation amount ψ_(ndop_code(ndm), ndm)(m) is represented by “phaserotator PROT #[ndop_code(ndm), ndm].” Likewise, transmission antenna 109that radiates the output of phase rotator PROT #[ndop_code(ndm), ndm]into a space is represented by “transmission antenna Tx#[ndop_code(ndm), ndm].” Here, ndm=1, . . . , N_(DM), andndop_code(ndm)=1, . . . , N_(DOP_CODE)(ndm).

For example, a description will be given of a case where number Nt oftransmission antennas used for multiplexing transmission is 3, numberN_(DM) of Doppler multiplexing is 2, number N_(CM) of code multiplexingis 2, orthogonal code sequences Code₁={1, 1} and Code₂={1, −1} with codelength Loc=2 are set, and numbers N_(DOP_CODE)(1) and N_(DOP_CODE)(2) ofcoded Doppler multiplexing are 1 and 2, respectively. In this case,coded Doppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), andψ_(2, 2)(m) are outputted from encoder 107 to phase rotators 108 in eachtransmission period.

For example, phase rotator PROT #[1, 1] applies, in each transmissionperiod, phase rotation amount ψ_(1, 1)(m) given by following Expression33 to a chirp signal generated by radar transmission signal generator101 in each transmission period. The output of phase rotator PROT #[1,1] is outputted from transmission antenna Tx #[1, 1]. Here, cp(t)denotes a chirp signal for each transmission period.

[33]

exp[jψ _(1,1)(1)]cp(t),exp[jψ _(1,1)(2)]cp(t),exp[jψ _(1,1)(3)]cp(t), .. . ,exp[jψ _(1,1)(Nc)]cp(t)   (Expression 33)

Likewise, phase rotator PROT #[1, 2] applies, in each transmissionperiod, phase rotation amount ψ_(1, 2)(m) given by following Expression34 to a chirp signal generated by radar transmission signal generator101 in each transmission period. The output of phase rotator PROT #[1,2] is outputted from transmission antenna Tx #[1, 2].

[34]

exp[jψ _(1,2)(1)]cp(t),exp[jψ _(1,2)(2)]cp(t),exp[jψ _(1,2)(3)]cp(t), .. . ,exp[jψ _(1,2)(Nc)]cp(t)   (Expression 34)

Likewise, phase rotator PROT #[2, 2] applies, in each transmissionperiod, phase rotation amount ψ_(2, 2)(m) given by following Expression35 to a chirp signal generated by radar transmission signal generator101 in each transmission period. The output of phase rotator PROT #[2,2] is outputted from transmission antenna Tx #[2, 2].

[35]

exp[jψ _(2,2)(1)]cp(t),exp[jψ _(2,2)(2)]cp(t),exp[jψ _(2,2)(3)]cp(t), .. . ,exp[jψ _(2,2)(Nc)]cp(t)   (Expression 35)

The foregoing description has been given of an example of setting ofcoded Doppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m).

Further, in the present embodiment, the arrangement of transmissionantennas 109 and the assignment of the coded Doppler phase rotationamounts are, for example, associated with each other as described below.This association allows radar apparatus 10 to utilize, in radarprocessing, the transmission antennas such that the number thereof ismade greater than the number of transmission antennas 109 formultiplexing transmission.

For example, at least a pair of adjacent transmission antennas 109transmit radar transmission signals using the same Doppler multiplexing(e.g., the same Doppler shift amount). For example, adjacentN_(DOP_CODE)(ndm__(BF)) transmission antennas 109 include transmissionantenna Tx #[1, ndm__(BF)], transmission antenna Tx #[2, ndm__(BF)], . .. , and transmission antenna Tx #[N_(DOP_CODE)(ndm__(BF)), ndm__(BF)] towhich phase rotator PROT #[1, ndm__(BF)], phase rotator PROT #[2,ndm__(BF)], . . . , and phase rotator PROT #[N_(DOP_CODE)(ndm__(BF)),ndm__(BF)] are assigned. Here, ndm__(BF) is any value of 1, . . . , andN_(DM) and is 1<N_(DOP_CODE)(ndm__(BF))≤N_(CM). In other words, among aplurality of combinations of Doppler shift amounts DOP_(ndm) and theorthogonal code sequences, the Doppler shift amount is the same (e.g.,ndm=ndm__(BF)) between combinations associated respectively withadjacent transmission antennas 109 of the plurality of transmissionantennas 109.

For example, one or more combinations may be included that satisfy theaforementioned association between the assignment of the coded Dopplerphase rotation amounts and the arrangement of transmission antennas 109.

By way of example, a description will be given of a case where number Ntof transmission antennas used for multiplexing transmission is 3, numberN_(DM) of Doppler multiplexing is 2, number N_(CM) of code multiplexingis 2, orthogonal code sequences Code₁={1, 1} and Code₂={1, −1} with codelength Loc=2 are used, and numbers N_(DOP_CODE)(1) and N_(DOP_CODE)(2)of coded Doppler multiplexing are 2 and 1, respectively. Note thatnumber N_(BF) of beam transmission antennas is set to 1, and ndm__(BF)=1is used as an index of a Doppler multiplexed signal used for the beamtransmission antenna.

In FIG. 12, for example, horizontally adjacent Nt=3 transmissionantennas 109 are transmission antenna Tx #[1, 1], transmission antennaTx #[2, 1], and transmission antenna Tx #[1, 2] from the left. In FIG.12, left two (=N_(DOP_CODE)(1)) adjacent transmission antennas Tx #[1,1] and Tx #[2, 1] (first sub-array antenna) transmit radar transmissionsignals using the same Doppler multiplexing (Doppler shift amount=DOP₁).

By way of another example, a description will be given of a case wherenumber Nt of transmission antennas used for multiplexing transmission is4, number N_(DM) of Doppler multiplexing is 2, number N_(CM) of codemultiplexing is 2, orthogonal code sequences Code₁={1, 1} and Code₂={1,−1} with code length Loc=2 are used, and numbers N_(DOP_CODE)(1) andN_(DOP_CODE)(2) of coded Doppler multiplexing are 2 and 2. Note thatnumber N_(BF) of beam transmission antennas is set to 2, andndm__(BF1)=1 and ndm__(BF2)=2 are used as indices of Doppler multiplexedsignals used for the beam transmission antennas.

In FIG. 13, for example, horizontally adjacent Nt=4 transmissionantennas 109 are transmission antenna Tx #[1, 1], transmission antennaTx #[2, 1], transmission antenna Tx #[1, 2], and transmission antenna Tx#[2, 2] from the left. In FIG. 13, left two (=N_(DOP_CODE)(1)) adjacenttransmission antennas Tx #[1, 1] and Tx #[2, 1] (first sub-arrayantenna) transmit radar transmission signals using the same Dopplermultiplexing (Doppler shift amount=DOP₁). Further, right two(=N_(DOP_CODE)(2)) adjacent transmission antennas Tx #[1, 2] and Tx #[2,2] (second sub-array antenna) transmit radar transmission signals usingthe same Doppler multiplexing (Doppler shift amount=DOP₂).

As is understood from these, at least one pair of adjacent transmissionantennas 109 transmit radar transmission signals using the same Dopplermultiplexing. In other words, at least one pair of adjacent transmissionantennas 109 perform code multiplexing transmission using the sameDoppler multiplexing.

Here, reception signals for each transmission period that correspond tothe radar transmission signals on which the code multiplexingtransmission is performed using the same Doppler multiplexing can beregarded as reception signals corresponding to orthogonal beamtransmission by a plurality of transmission antennas 109. For example,the transmission by at least one pair of adjacent transmission antennas109 described above is equivalent to orthogonal beam transmission by asub-array composed of such adjacent transmission antennas 109. When aradar transmission signal is transmitted from the at least one pair ofadjacent transmission antennas 109 described above, for example, at anequal power, such transmission may be treated as transmission by a newtransmission antenna (hereinafter, referred to as a “beam transmissionantenna”) for which a midpoint position between adjacent transmissionantennas 109 serves as the phase center of the sub-array (details willbe described later in conjunction with the reception processing). Notethat, when the radar transmission signal is not transmitted at an equalpower from transmission antennas 109 constituting the beam transmissionantenna, transmission at a position dependent on the ratio oftransmission powers of respective transmission antennas 109 constitutingthe beam transmission antenna (the position of the center of gravity ofthe transmission powers from the respective transmission antennas) thatserves as the phase center of the sub-array can be treated astransmission by the beam transmission antenna.

The transmission method as described above allows radar apparatus 10 toutilize the transmission antennas such that the number thereof is madegreater than number Nt of transmission antennas for multiplexingtransmission.

Note that, the example in which transmission antennas 109 arehorizontally arranged has been described with reference to FIGS. 12 and13, the arrangement method of transmission antennas 109 is not limitedthereto. For example, transmission antennas 109 may be verticallyarranged, or may be arranged in a horizontal and vertical plane.Further, antennas constituting transmission antennas 109 may be composedof a plurality of horizontally arranged sub-array elements, a pluralityof vertically arranged sub-array elements, or a plurality of sub-arrayelements arranged in a horizontal and vertical plane. In addition, theantennas illustrated in FIGS. 12 and 13 may be a part of a plurality ofantennas that radar apparatus 10 includes.

As is understood, in the present embodiment, each of a plurality oftransmission antennas 109 is associated with the combination (in otherwords, assignment) of Doppler shift amount DOP_(ndm) and orthogonal codesequence Code_(ncm) such that at least one of Doppler shift amountDOP_(ndm) and orthogonal code sequence Code_(ncm) differs fromcombination to combination.

Further, in the present embodiment, when numbers N_(DOP_CODE)(ndm) ofcoded Doppler multiplexing for the Doppler multiplexed signals are setnon-uniformly, the numbers of multiplexing by orthogonal code sequencesCode_(ncm) (in other words, numbers N_(DOP_CODE)(ndm) of coded Dopplermultiplexing) corresponding respectively to Doppler shift amountsDOP_(ndm) may be different among the combinations of Doppler shiftamounts DOP_(ndm) and orthogonal code sequences Code_(ncm). By way ofexample, as illustrated in FIG. 3, Nt transmission antennas 109 may atleast include a plurality of (e.g., two) transmission antennas 109 fromwhich transmission signals that are code-multiplexed using differentorthogonal code sequences are transmitted, and at least one transmissionantenna 109 from which a transmission signal that is notcode-multiplexed is transmitted. In other words, radar transmissionsignals transmitted from radar transmitter 100 include at least a codedDoppler multiplexed signal for which number N_(DOP_CODE)(ndm) of codedDoppler multiplexing is set to number N_(CM) of codes, and a codedDoppler multiplexed signal for which number N_(DOP_CODE)(ndm) of codedDoppler multiplexing is set smaller than number N_(CM) of codes.

Further, in the present embodiment, when numbers N_(DOP_CODE)(ndm) ofcoded Doppler multiplexing for the Doppler multiplexed signals are setuniformly, the numbers of multiplexing by orthogonal code sequencesCode_(ncm) (in other words, numbers N_(DOP_CODE)(ndm) of coded Dopplermultiplexing) corresponding respectively to Doppler shift amountsDOP_(ndm) may be the same among the combinations of Doppler shiftamounts DOP_(ndm) and orthogonal code sequences Code_(ncm).

[Configuration of Radar Receiver 200]

In FIG. 1, radar receiver 200 includes Na reception antennas 202, whichconstitute an array antenna. Radar receiver 200 further includes Naantenna system processors 201-1 to 201-Na, constant false alarm rate(CFAR) section 211, coded Doppler demultiplexer 212, peak extractor 213,and direction estimator 214.

Each of reception antennas 202 receives a reflected wave signal that isa radar transmission signal reflected from a target object (target), andoutputs the received reflected wave signal to the corresponding one ofantenna system processors 201 as a reception signal.

Each of antenna system processors 201 includes reception radio 203 andsignal processor 206.

Reception radio 203 includes mixer 204 and low pass filter (LPF) 205.Reception radio 203 mixes, at mixer 204, a chirp signal inputted fromradar transmission signal generator 101, which is a transmission signal,with the received reflected wave signal, and passes the resulting mixedsignal through LPF 205. As a result, a beat signal having a frequencycorresponding to the delay time of the reflected wave signal isacquired. For example, as illustrated in FIG. 14, the differencefrequency between the frequency of a transmission chirp signal(transmission frequency-modulated wave) and the frequency of a receptionchirp signal (reception frequency-modulated wave) is obtained as a beatfrequency.

In each antenna system processor 201-z (where z is any of 1 to Na),signal processor 206 includes analog-to-digital (AD) converter 207, beatfrequency analyzer 208, output switch 209, and Doppler analyzers 210.

The signal (for example, beat signal) outputted from LPF 205 isconverted into discretely sampled data by AD converter 207 in signalprocessor 206.

Beat frequency analyzer 208 performs, in each transmission period Tr,FFT processing on N_(data) pieces of discretely sampled data obtained ina defined time range (range gate). Signal processor 206 thus outputs afrequency spectrum in which a peak appears at a beat frequency dependenton the delay time of the reflected wave signal (radar reflected wave).In the FFT processing, for example, beat frequency analyzer 208 mayperform multiplication by a window function coefficient such as the Hanwindow or the Hamming window. The use of the window function coefficientcan suppress sidelobes around the beat frequency peak.

Here, a beat frequency response obtained from the m-th chirp pulsetransmission, which is outputted from beat frequency analyzer 208 inz-th signal processor 206, is represented by RFT_(z)(f_(b), m). Here,f_(b) denotes the beat frequency index and corresponds to an FFT index(bin number). For example, f_(b)=0, . . . , N_(data)/2−1, z=1, . . . ,Na, and m=1, . . . , Nc. A beat frequency having smaller beat frequencyindex f_(b) indicates a shorter delay time of the reflected wave signal(in other words, a shorter distance to the target object).

In addition, beat frequency index f_(b) may be converted into distanceinformation R(f_(b)) using following Expression 36. Thus, in thefollowing, beat frequency index f_(b) is also referred to as “distanceindex f_(b).”

$\begin{matrix}\lbrack 36\rbrack & \; \\{{R\left( f_{b} \right)} = {\frac{C_{0}}{2B_{w}}f_{b}}} & \left( {{Expression}\mspace{14mu} 36} \right)\end{matrix}$

Here, B_(w) denotes a frequency-modulation bandwidth within the rangegate for a chirp signal, and CO denotes the speed of light.

Output switch 209 performs selective switching to output the output ofbeat frequency analyzer 208 for each transmission period to OC_INDEX-thDoppler analyzer 210 among Loc Doppler analyzers 210 based on orthogonalcode element index OC_INDEX inputted from encoder 107 of phase rotationamount setter 105. In other words, output switch 209 selects OC_INDEX-thDoppler analyzer 210 given by Expression 11 for m-th transmission periodTr.

Signal processor 206 includes Loc Doppler analyzers 210-1 to 210-Loc.For example, data is inputted by output switch 209 to noc-th Doppleranalyzer 210 in each of Loc transmission periods (Loc×Tr). Accordingly,noc-th Doppler analyzer 210 performs Doppler analysis for each distanceindex f_(b) using data of Ncode transmission periods among Nctransmission periods (for example, using beat frequency responseRFT_(z)(f_(b), m) inputted from beat frequency analyzer 208). Here, nocdenotes the index of a code element, and noc=1, . . . , Loc.

For example, when Ncode is a power of 2, FFT processing is applicable inthe Doppler analysis. In this case, the FFT size is Ncode, and a maximumDoppler frequency that is derived from the sampling theorem and in whichno aliasing occurs is ±1/(2Loc×Tr). Further, the Doppler frequencyinterval of Doppler frequency index f_(s) is 1/(Ncode×Loc×Tr), and therange of Doppler frequency index f_(s) is given by f_(s)=Ncode/2, . . ., 0, . . . , Ncode/2−1.

The following description will be given of a case where Ncode is a powerof 2, as an example. Note that, when Ncode is not a power of 2,zero-padded data is included, for example, to allow FFT processing to beperformed, with the data size (FFT size) being equal to a power of 2. Inthe FFT processing, Doppler analyzer 210 may perform multiplication by awindow function coefficient such as the Han window or the Hammingwindow. The application of a window function can suppress sidelobesaround the Doppler frequency peak.

For example, output VFT_(z) ^(noc)(f_(b), f_(s)) of Doppler analyzer 210of z-th signal processor 206 is given by following Expression 37. Here,j is the imaginary unit and z=1 to Na.

$\begin{matrix}{\mspace{79mu}\lbrack 37\rbrack} & \; \\{{{VF}{T_{z}^{noc}\left( {f_{b},f_{s}} \right)}} = {\sum\limits_{s = 0}^{N_{code} - 1}{RF{T_{z}\left( {f_{b},{{L_{OC} \times s} + {noc}}} \right)}{\exp\left\lbrack {{- j}\frac{2\pi sf_{s}}{N_{code}}} \right\rbrack}}}} & \left( {{Expression}\mspace{14mu} 37} \right)\end{matrix}$

The processing by the constituent sections of signal processor 206 hasbeen described above.

In FIG. 1, CFAR section 211 performs CFAR processing (in other words,adaptive threshold judgement) using the outputs of Loc Doppler analyzers210 in each of the first to Na-th signal processors 206 and extractsdistance indices f_(b_cfar) and Doppler frequency indices f_(s_cfar)that provide peak signals.

For example, CFAR section 211 performs power addition of outputs VFT_(z)^(noc)(f_(b), f_(s)) of Doppler analyzers 210 in first to Na-th signalprocessors 206, for example, as given by following Expression 38, so asto perform two-dimensional CFAR processing in two dimensions formed bythe distance axis and the Doppler frequency axis (corresponding to therelative velocity) or CFAR processing using one-dimensional CFARprocessing in combination. For example, processing disclosed in NPL 2may be applied as the two-dimensional CFAR processing or the CFARprocessing using one-dimensional CFAR processing in combination.

$\begin{matrix}\lbrack 38\rbrack & \; \\{{{PowerFT}\left( {f_{b},f_{s}} \right)} = {\sum\limits_{z = 1}^{N_{a}}{\sum\limits_{{noc} = 1}^{L_{oc}}{{{VF}{T_{z}^{noc}\left( {f_{b},f_{s}} \right)}}}^{2}}}} & \left( {{Expression}\mspace{14mu} 38} \right)\end{matrix}$

CFAR section 211 adaptively sets a threshold and outputs, to codedDoppler demultiplexer 212, distance index f_(b_cfar) and Dopplerfrequency index f_(s_cfar) that provide a received power greater thanthe threshold, and received-power information PowerFT(f_(b_cfar),f_(s_cfar)).

For example, when phase rotation amount φ_(ndm) for applying Dopplershift amount DOP_(ndm) is determined using Expression 5, the intervalsbetween the Doppler shift amounts in the Doppler frequency domain, whichare outputted from Doppler analyzers 210, are equal intervals, andΔFD=Ncode/N_(DM) when intervals ΔFD of the Doppler shift amounts arerepresented by the intervals of the Doppler frequency indices.Accordingly, in the outputs of Doppler analyzers 210, a peak is detectedfor each Doppler-shift multiplexed signal at an interval of ΔFD in theDoppler frequency domain. Note that, when phase rotation amount φ_(ndm)is determined using Expression 5, ΔFD may sometimes not be an integerdepending on Ncode and N_(DM). In this case, Expression 59 describedbelow may be used to obtain ΔFD having an integer value. The followingdescribes a reception processing operation using ΔFD having an integervalue.

Part (a) in FIG. 15 illustrates an example of the outputs of Doppleranalyzers 210 for the distances over which reflected waves from threetargets exist in a case where N_(DM)=2. For example, as illustrated at(a) in FIG. 15, when reflected waves from the three targets are observedat Doppler frequency indices f1, f2, and f3, the reflected waves arealso observed at respective Doppler frequency indices spaced from f1, f2and f3 by the interval of ΔFD (for example, f1−ΔFD, f2−ΔFD, andf3−ΔFD+Ncode).

Accordingly, CFAR section 211 may perform, as given by followingExpression 39, power addition (referred to as, for example, “Dopplerdomain compression”) with respect to the outputs of Doppler analyzers210 while adjusting peak positions of Doppler-shift multiplexed signalsto respective ranges resulting from division by the range of intervalΔFD of the Doppler shift amounts. Subsequently, CFAR section 211 mayperform CFAR processing (referred to as, for example, “Doppler domaincompression CFAR processing”). Here, f_(s_comp)=ΔFD/2, . . . , ΔFD/2−1.For example, when ΔFD=Ncode/N_(DM), then f_(s_comp)=Ncode/(2N_(DM)), . .. , Ncode/(2N_(DM))−1.

$\begin{matrix}{\mspace{79mu}\lbrack 39\rbrack} & \; \\{{{Powe}rF{T_{comp}\left( {f_{b},f_{s\_ comp}} \right)}} = {\sum\limits_{{nfd} = 1}^{N_{DM}}{{PowerFT}\left( {\quad{f_{b},\left. \quad{f_{s\_ comp} + \ {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM}}{2} \right)} - 1} \right) \times \Delta FD}} \right)}} \right.}}} & \left( {{Expression}\mspace{14mu} 39} \right)\end{matrix}$

However, in Expression 39, in the case of

$\begin{matrix}\lbrack 40\rbrack & \; \\{{{f_{s\_ comp} + \ {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM}}{2} \right)} - 1} \right) \times \Delta FD}} < {{- N}c{ode}\text{/}2}},} & \;\end{matrix}$

the Doppler frequency index to which Ncode is added is used.

Likewise, in Expression 39, in the case of

$\begin{matrix}\lbrack 41\rbrack & \; \\{{{f_{s_{-}comp} + {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM}}{2} \right)} - 1} \right) \times \Delta FD}} > {\frac{Ncode}{2} - 1}},} & \;\end{matrix}$

the Doppler frequency index from which Ncode is further subtracted isused.

It is thus possible to compress the Doppler frequency range for the CFARprocessing to 1/N_(DM) to reduce the amount of CFAR processing and tosimplify the circuit configuration. In addition, CFAR section 211 iscapable of power addition for N_(DM) Doppler-shift multiplexed signals,to improve a signal to noise ratio (SNR) by about (N_(DM))^(1/2). As aresult, the radar sensing performance of radar apparatus 10 can beimproved.

Part (b) in FIG. 15 illustrates an example of outputs that are theoutputs of Doppler analyzers 210 illustrated at (a) in FIG. 15 to whichthe Doppler domain compression processing given by Expression 39 isapplied. As illustrated at (b) in FIG. 15, in a case where N_(DM)=2,CFAR section 211 adds together the power component for Doppler frequencyindex f1 and the power component for f1−ΔFD through the Doppler domaincompression processing and outputs the result. Likewise, as illustratedat (b) in FIG. 15, CFAR section 211 adds together the power component ofDoppler frequency index f2 and the power component of f2−ΔFD and outputsthe result. Further, regarding the power component of Doppler frequencyindex f3, f3−ΔFD is smaller than −Ncode/2. Thus, CFAR section 211 addstogether the power component of Doppler frequency index f3 and the powercomponent of f3−ΔFD+Ncode (f3+ΔFD when N_(DM)=2, for example) andoutputs the result.

As a result of the Doppler domain compression, the range of Dopplerfrequency indices f_(s_comp) in the Doppler frequency domain is reducedto the range of from ΔFD/2 through ΔFD/2−1 (when ΔFD=Ncode/N_(DM), therange is from Ncode/(2N_(DM)) through Ncode/(2N_(DM))−1) and the rangeof the CFAR processing is compressed, resulting in reduction of thecomputation amount of CFAR processing. In addition, in FIG. 15, forexample, because of power addition for the reflected waves from thethree targets, the SNR of the signal components is improved. Note thatthe power of noise components is also combined, and thus, improvement inSNR is, for example, about (N_(DM))^(1/2).

For example, CFAR section 211, which uses the Doppler domain compressionCFAR processing, adaptively sets a threshold and outputs, to codedDoppler demultiplexer 212, distance index f_(b_cfar) and Dopplerfrequency index f_(s_comp_cfar) that provide a received power greaterthan the threshold, and received-power information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil(N_(DM)/2)−1)×ΔFD (where nfd=1, . . . ,N_(DM))) for Doppler frequency indices(f_(s_comp_cfar)+(nfd−ceil(N_(DM)/2)−1)×ΔFD) of N_(DM) Dopplermultiplexed signals.

In addition, CFAR section 211 outputs, for example, distance indexf_(b_cfar) and Doppler frequency index f_(s_comp_cfar) to peak extractor213.

Note that phase rotation amount φ_(ndm) for applying Doppler shiftamount DOP_(ndm) is not limited to that given by Expression 5. CFARsection 211 can apply the Doppler domain compression CFAR processing,for example, if phase rotation amounts φ_(ndm) of Doppler-shiftmultiplexed signals are such that peaks are detected at constantintervals in the Doppler frequency domain in the outputs from Doppleranalyzers 210.

For example, when Δf_(MinInterval)=1/(T_(r)(N_(DM)+N_(int))LOC) is setusing the equal-interval Doppler shift amount setting, phase rotationamount φ_(ndm) is set according to Expression 6, and the Doppler-shiftmultiplexed signals are detected as peaks at the intervals ofΔFD=Ncode/(N_(DM)+N_(int)) in the Doppler frequency domain in theoutputs from the Doppler analyzers 210. Also in such a case, CFARsection 211 can apply the Doppler domain compression CFAR processing.

Next, an example of the operation of coded Doppler demultiplexer 212illustrated in FIG. 1 will be described. The following describes anexample of processing performed by coded Doppler demultiplexer 212 whenCFAR section 211 uses the Doppler domain compression CFAR processing.

Based on the outputs of CFAR section 211 (e.g., distance indicesf_(b_cfar), Doppler frequency indices f_(s_comp_cfar), andreceived-power information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil(N_(DM)/2)−1)×ΔFD) for Doppler frequencyindices (f_(s_comp_cfar)+(nfd−ceil(N_(DM)/2)−1)×ΔFD (where nfd=1, . . ., N_(DM))) of N_(DM) Doppler multiplexed signals), coded Dopplerdemultiplexer 212 separates, using the outputs of Doppler analyzers 210,coded-Doppler multiplexed signals transmitted, and distinguishes (inother words, also referred to as “judges” or “identifies”) transmissionantennas 109 and Doppler frequencies (in other words, Doppler velocitiesor relative velocities).

As described above, when the equal-interval Doppler shift amount settingincluding the maximum equal-interval Doppler shift amount setting isused, encoder 107 in phase rotation amount setter 105 does not set allof N_(DM) numbers N_(DOP_CODE)(1), N_(DOP_CODE)(2), . . . , andN_(DOP_CODE)(N_(DM)) of coded Doppler multiplexing to N_(CM), forexample, but may set at least one of the numbers of coded Dopplermultiplexing to a value smaller than N_(CM). For example, coded Dopplerdemultiplexer 212 performs (1) code separation processing to detect acoded Doppler multiplexed signal for which the number of coded Dopplermultiplexing is set smaller than N_(CM) (in other words, detect anunused coded Doppler multiplexed signal that is not used formultiplexing transmission), to perform aliasing judgement. Thereafter,coded Doppler demultiplexer 212 performs (2) Doppler code separationprocessing on coded Doppler multiplexed signals used for multiplexingtransmission based on an aliasing judgement result.

Processing (1) and processing (2) by coded Doppler demultiplexer 212described above will be described below.

<(1) Aliasing Judgement Processing (Detection Processing of DetectingUnused Coded Doppler Multiplexed Signal)>

Coded Doppler demultiplexer 212 performs the Doppler aliasing judgementprocessing, for example, on the assumption that the Doppler range of atarget is ±1/(2Tr).

Here, each of Doppler analyzers 210 applies the FFT processing to eachcode element, for example, when Ncode is a power value of 2, and thusperforms the FFT processing in periods of (Loc×Tr) using the output frombeat frequency analyzer 208. Thus, for Doppler analyzer 210, the Dopplerrange in which no aliasing occurs according to the sampling theorem is±1/(2Loc×Tr). Since Doppler multiplexing is further performed on thisDoppler range of ±1/(2Loc×Tr) by using number N_(DM) of Dopplermultiplexing, coded Doppler demultiplexer 212 performs the aliasingjudgement processing, assuming the Doppler range of ±1/(2Tr) resultingfrom multiplication, by Loc×N_(DM), of the Doppler range of±1/(2Loc×N_(DM)×Tr) in which aliasing due to Doppler multiplexing doesnot occur.

Here, by way of example, a description will be given of a case where Ntis 3, number N_(DM) of Doppler multiplexing is 2, and number N_(CM) ofcode multiplexing is 2. Here, phase rotation amount φ_(ndm) for applyingDoppler shift amount DOP_(ndm) is assigned, for example, as given byExpression 5 that is based on the maximum equal-interval Doppler shiftamount setting. In this case, phase rotation amount φ₁ for applyingDoppler shift amount DOP₁ is 0, and, phase rotation amount φ₂ forapplying Doppler shift amount DOP₂ is 71 In addition, encoder 107 usestwo orthogonal codes Code₁={1, 1} and Code₂={1, −1} among Walsh-Hadamardcodes having code length Loc=2. Further, as illustrated at (a) in FIG.3, N_(DOP_CODE)(1)=2 and N_(DOP_CODE)(2)=1 are used.

In this case, with respect to the Doppler range of±1/(2Loc×N_(DM)×Tr)=±1/(8Tr) in which no aliasing due to the codedDoppler multiplexing occurs, coded Doppler demultiplexer 212 performsthe aliasing judgement processing assuming the Doppler range of ±1/(2Tr)resulting from multiplication of the Doppler range of±1/(2Loc×N_(DM)×Tr)=±1/(8Tr) by 4 (=Loc×N_(DM)).

Here, Doppler component VFT_(z) ^(noc)(f_(b_cfar), f_(s_comp_cfar)),which is the output of Doppler analyzer 210 corresponding to distanceindex f_(b_cfar) and Doppler frequency index f_(s_comp_cfar) extractedin CFAR section 211, may contain a Doppler component including aliasingas illustrated at (a) and (b) in FIG. 16, for example, in the DopplerRange of ±1/(2Tr).

For example, as illustrated at (a) in FIG. 16, there is a possibility offour (=Loc×N_(DM)) Doppler components of f_(s_comp_cfar)−Ncode/N_(DM),f_(s_comp_cfar), f_(s_comp_cfar)+Ncode/N_(DM), andf_(s_comp_cfar)+2Ncode/N_(DM) in the case of f_(s_comp_cfar)<0 withinthe Doppler range of ±1/(2Tr) (the Doppler components may also beexpressed using ΔFD=Ncode/N_(DM) as f_(s_comp_cfar)−ΔFD,f_(s_comp_cfar), f_(s_comp_cfar)+ΔFD, and f_(s_comp_cfar)+2ΔFD).

Further, for example, as illustrated at (b) in FIG. 16, there is apossibility of four (=Loc×N_(DM)) Doppler components off_(s_comp_cfar)−2Ncode/N_(DM), f_(s_comp_cfar)−Ncode/N_(DM),f_(s_comp_cfar), and f_(s_comp_cfar)+Ncode/N_(DM) in the case off_(s_comp_cfar)>0 within the Doppler range of ±1/(2Tr) (the Dopplercomponents may also be expressed using ΔFD=Ncode/N_(DM) asf_(s_comp_cfar)−2ΔFD, f_(s_comp_cfar)−ΔFD, f_(s_comp_cfar), andf_(s_comp_cfar)+ΔFD). These possible Doppler components (four(=Loc×N_(DM)) components) with respect to f_(s_comp_cfar) are called“Doppler component candidates” with respect to f_(s_comp_cfar). In thefollowing, Doppler regions in which such four (=Loc×N_(DM)) Dopplercomponent candidates are present are represented using index “D_(r)”indicating the Doppler aliasing range as illustrated in FIG. 16. D_(r)is an index indicating the Doppler aliasing range, and for example, aninteger value in a range of D_(r)∈{−ceil(Loc×N_(DM)/2),ceil(Loc×N_(DM)/2)−1} is used. In FIG. 16, D_(r)=2, . . . , 1. Notethat, the region where D_(r)=0 indicates the region where no Doppleraliasing occurs, and the regions where D_(r)≠0 indicate the regionswhere Doppler aliasing occurs. Further, the greater the absolute valueof D_(r), the more distant the Doppler region is from the Doppler regionindicated by D_(r)=0.

Coded Doppler demultiplexer 212 corrects phase changes corresponding tothe four (=Loc×N_(DM)) Doppler components including aliasing within theDoppler range of ±1/(2Tr) as illustrated in FIG. 16, and performs thecoded Doppler demultiplexing processing on the coded Doppler multiplexedsignal for which the number of coded Doppler multiplexing is set smallerthan N_(CM) (in other words, the unused coded Doppler multiplexedsignal).

Then, based on the received power of components obtained by the codedDoppler demultiplexing processing on the unused coded Dopplermultiplexed signal, coded Doppler demultiplexer 212 judges whether ornot each of the Doppler component candidates is a true Dopplercomponent.

For example, coded Doppler demultiplexer 212 may detect, among theDoppler component candidates with respect to f_(s_comp_cfar), a Dopplercomponent having the minimum received power obtained by the codedDoppler demultiplexing processing based on the unused coded Dopplermultiplexed signal, to judge that the detected Doppler component is thetrue Doppler component. In other words, coded Doppler demultiplexer 212may judge that those of the Doppler component candidates with respect tof_(s_comp_cfar) which have received powers different from the minimumreceived power are false Doppler components.

This aliasing judgement processing can resolve the ambiguity in theDoppler range of ±1/(2Tr). In addition, by this aliasing judgementprocessing, the range in which a Doppler frequency can be detectedwithout ambiguity can be extended to a range of from −1/(2Tr) to lessthan 1/(2Tr) as compared with the Doppler range of±1/(2Loc×N_(DM)×Tr)=±1/(8Tr) in which aliasing due to Dopplermultiplexing does not occur.

By the coded Doppler demultiplexing based on the unused coded Dopplermultiplexed signal, for example, the phase change of the true Dopplercomponent is corrected and the orthogonality between the coded Dopplermultiplexed signals used for multiplexing transmission and the unusedcoded Doppler multiplexed signal is maintained. Therefore, the codedDoppler multiplexed signal codes used for multiplexing transmission andthe unused coded Doppler multiplexed signal become uncorrelated, and thereceived power becomes as low as a noise level.

On the other hand, the phase changes of false Doppler components are,for example, erroneously corrected and the orthogonality between thecoded Doppler multiplexed signals used for multiplexing transmission andthe unused coded Doppler multiplexed signal is not maintained. Thus, acorrelated component (interference component) is caused between thecoded Doppler multiplexed signal codes used for the multiplexingtransmission and the unused coded Doppler multiplexed signal, and, forexample, a received power greater than a noise level may be detected.Therefore, as described above, coded Doppler demultiplexer 212 may judgethat the Doppler component having the minimum received power among theDoppler component candidates with respect to f_(s_comp_cfar) resultingfrom the coded Doppler demultiplexing based on the unused coded Dopplermultiplexed signal is the true Doppler component, and judge that theother Doppler components having the received powers different from theminimum received power are the false Doppler components.

For example, coded Doppler demultiplexer 212 corrects the phase changescorresponding to the Doppler components of the Doppler componentcandidates with respect to f_(s_comp_cfar) based on the outputs ofDoppler analyzers 210 in each antenna system processor 201, andcalculates, according to Expression 40, received powerP_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(r), nuc, nud) after the codeseparation using the unused coded Doppler multiplexed signal.

Here, nuc and nud represent an index of an orthogonal code serving asthe unused coded Doppler multiplexed signal and an index of the Dopplermultiplexed signal, respectively. For example, in the case of (b) ofFIG. 3, the unused coded Doppler multiplexed signal is indicated by across mark in the figure, is assigned the code of Code₂, and is assignedthe Doppler shift amount of DOP₁. Accordingly, indices nuc and nud ofthe orthogonal code to which the unused coded Doppler multiplexed signalis assigned are 2 and 1, respectively.

In the following, a pair of the index of the orthogonal code and theindex of the Doppler multiplexed signal that are used for the codedDoppler multiplexed signal is described as “DCI (index of orthogonalcode, index of Doppler multiplexed signal).” DCI (nuc, nud) represents,for example, the index of an orthogonal code to which an unused codedDoppler multiplexed signal is assigned and the index of a Dopplermultiplexed signal. For example, in the case of (b) in FIG. 3, theunused coded Doppler multiplexed signal is assigned to DCI (2, 1).Similarly, for example, in the case of (a) in FIG. 5, the unused codedDoppler multiplexed signal is assigned to DCI (2, 2) and DCI (2, 3).Further, for example, in the case of (c) in FIG. 6, the unused codedDoppler multiplexed signal is assigned to DCI (1, 2) and DCI (2, 3).

$\begin{matrix}{\mspace{79mu}\lbrack 42\rbrack} & \; \\{{P_{DAR}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r},{nuc},{nud}} \right)} = {\sum\limits_{z = 1}^{N_{a}}{{Y_{z}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r},{nuc},{nud}} \right)}}^{2}}} & \left( {{Expression}\mspace{14mu} 40} \right)\end{matrix}$

Here, Y_(z)(f_(b_cfar), f_(s_comp_cfar), D_(r), nuc, nud) is a receivedsignal obtained by correction of the phase changes corresponding to theDoppler components of Doppler component candidates with respect tof_(s_comp_cfar) and separation of the unused coded Doppler multiplexedsignal to which DCI (nuc, nud) is assigned. The correction andseparation are based on the outputs of Doppler analyzers 210 in z-thantenna system processor 201 as given by following Expression 41:

$\begin{matrix}{\mspace{85mu}\lbrack 43\rbrack} & \; \\{{Y_{z}\left( {f_{b_{cfar}},f_{s_{{comp}_{cfar}}},D_{r},{nuc},{nud}} \right)} = {{Code}_{nuc}^{*}{\left\{ {{\alpha\left( {f_{{s\_ comp}{\_ cfar}},D_{r}} \right)} \otimes \left( {f_{b\_ cfar},f_{{s\_ comp}{\_ cfar}},D_{r},{nud}} \right)} \right\}.}}} & \left( {{Expression}\mspace{14mu} 41} \right)\end{matrix}$

In Expressions 40 and 41, in order to separate the unused coded Dopplermultiplexed signal to which DCI (nuc, nud) is assigned, the receivedpowers after code separation using unused orthogonal code Code_(nuc) arecalculated with respect to outputs VFTALL_(z)(f_(b_cfar),f_(s_comp_cfar), D_(r), nud) of Doppler analyzers 210 in z-th antennasystem processor 201, and the sum of such powers is calculated withrespect to all the antenna system processors 201. Thus, it is possibleto increase the aliasing judgement accuracy even when the receivedsignal level is low. However, instead of Expression 40, the receivedpowers after code separation using the unused coded Doppler multiplexedsignal may be calculated with respect to the outputs of Doppleranalyzers 210 in some of antenna system processors 201. Even in thiscase, it is possible to reduce the arithmetic processing amount whilemaintaining the aliasing judgement accuracy, for example, in a rangewhere the received signal level is sufficiently high.

Note that, in Expressions 40 and 41, D_(r) is an index indicating theDoppler aliasing range, and takes an integer value in a range ofD_(r)∈{−ceil(Loc×N_(DM)/2), . . . , ceil(Loc×N_(DM)/2)−1}, for example.

Further, in Expression 41,

operator “⊗”  [44]

represents a product between each pair of elements of vectors having thesame number of elements. For example, for n-th order vectors A=[a₁, . .. , a_(n)] and B=[b₁, . . . , b_(n)], the product between each pair ofelements is expressed by following Expression 42:

[45]

A⊗B=[a ₁ , . . . ,a _(n)]⊗[b ₁ , . . . ,b _(n)]=[a ₁ b ₁ , . . . ,a _(n)b _(n)]  (Expression 42).

Moreover, in Expression 41, superscript T represents vectortransposition, and superscript * (asterisk) represents a complexconjugate operator.

In Expression 41, α(f_(s_comp_cfar), D_(r)) represents “Doppler phasecorrection vector.” Doppler phase correction vector α(f_(s_comp_cfar),D_(r)) corrects the Doppler phase rotation caused by a time differencebetween Doppler analyses of Loc Doppler analyzers 210 within Doppleraliasing range D_(r) when Doppler frequency index f_(s_comp_cfar)extracted in CFAR section 211 is in an output range (in other words,Doppler range) of Doppler analyzers 210 that does not include Doppleraliasing, for example.

For example, Doppler phase correction vector α(f_(s_comp_cfar), D_(r))is expressed by following Expression 43. For example, Doppler phasecorrection vector α(f_(s_comp_cfar), D_(r)) as given by Expression 43 isa vector having, as an element, a Doppler phase correction factor. TheDoppler phase correction factor corrects phase rotations of Dopplercomponents having Doppler frequency indices f_(s_comp_cfar) and beingwithin Doppler aliasing range D_(r). The phase rotations are caused bythe time lags of Tr, 2Tr, . . . , (Loc−1)Tr of the outputs of fromoutput VFT_(z) ²(f_(b_cfar), f_(s_comp_cfar)) of second Doppler analyzer210 to output VFT_(z) ^(Loc)(f_(b_cfar), f_(s_comp_cfar)) of Loc-thDoppler analyzer 210, for example, with reference to the Doppleranalysis time for analysis on output VFT_(z) ¹(f_(b_cfar),f_(s_comp_cfar)) of first Doppler analyzer 210. Note that, the term“D_(r)N_(code)/N_(DM)” in Expression 43 can also be expressed as“D_(r)ΔFD” using ΔFD=Ncode/N_(DM). Therefore, the expression isapplicable in the other cases than the case of ΔFD=Ncode/N_(DM).

$\begin{matrix}{\mspace{79mu}\lbrack 46\rbrack} & \; \\{{\alpha\left( {f_{{s\_ comp}{\_ cfar}},D_{r}} \right)} = {\quad\left\lbrack {1,{\exp\left( {{- \frac{j\; 2\;{\pi\begin{pmatrix}{f_{{s\_ comp}{\_ cfar}} +} \\\frac{D_{r}N_{code}}{N_{DM}}\end{pmatrix}}}{N_{code}}} \times \frac{1}{L_{oc}}} \right)},\left. \quad{\ldots\mspace{14mu},{\exp\left( {{- \frac{j\; 2\;{\pi\begin{pmatrix}{f_{{s\_ comp}{\_ cfar}} +} \\\frac{D_{r}N_{code}}{N_{DM}}\end{pmatrix}}}{N_{code}}} \times \frac{\left( {L_{oc} - 1} \right)}{L_{oc}}} \right)}} \right\rbrack^{T}} \right.}} & \left( {{Expression}\mspace{14mu} 43} \right)\end{matrix}$

Such phase correction by Doppler phase correction vectorα(f_(s_comp_cfar), D_(r)) corresponds to correction of phase changescorresponding to Doppler components of the Doppler component candidateswith respect to f_(s_comp_cfar).

Further, for example, as given by following Expression 44,VFTALL_(z)(f_(b_cfar), f_(s_comp_cfar), D_(r), nud) in Expression 41represents, in a vector format, components of an unused coded Dopplermultiplexed signal to which DCI (nuc, nud) is assigned. The componentsare extracted within Doppler aliasing range D_(r). The unused codedDoppler multiplexed signal corresponds to distance index f_(b_cfar) andDoppler frequency index f_(s_comp_cfar) extracted in CFAR section 211among outputs VFT_(z) ^(noc)(f_(b), f_(s)) of Loc Doppler analyzers 210in z-th antenna system processor 201. Here, noc=1, . . . , Loc and noctakes an integer value in the range of D_(r)={−ceil(Loc×N_(DM)/2), . . ., ceil(Loc×N_(DM)/2)−1}.

$\begin{matrix}{\mspace{79mu}\lbrack 47\rbrack} & \; \\{{{VFTA}L{L_{z}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r},{nud}} \right)}} = {\quad\left\lbrack {{{VFT}_{z}^{1}\left( {f_{b_{c{far}}},{f_{s_{{comp}_{cfar}}} + \frac{\begin{matrix}{N_{code}F_{R}} \\\left( {D_{r},{nud}} \right)\end{matrix}}{N_{DM}}}} \right)}\mspace{14mu}\ldots\mspace{14mu}\left. \quad{{VFT}_{z}^{L_{oc}}\left( {f_{b_{c{far}}},{f_{s_{{comp}_{cfar}}} + \frac{\begin{matrix}{N_{code}F_{R}} \\\left( {D_{r},{nud}} \right)\end{matrix}}{N_{DM}}}} \right)} \right\rbrack^{T}} \right.}} & \left( {{Expression}\mspace{14mu} 44} \right)\end{matrix}$

In Expression 44, N_(code)F_(R)(D_(r), nud)/N_(DM) represents an offsetvalue of the Doppler index of the nud-th Doppler multiplexed signal withrespect to f_(s_comp_cfar) within Doppler aliasing range D_(r). Notethat, the term “N_(code)F_(R)(D_(r), nud)/N_(DM)” in Expression 44 canalso be expressed as F_(R)(D_(r), nud)ΔFD using ΔFD=Ncode/N_(DM).Therefore, the expression is applicable in the other cases than the caseof ΔFD=Ncode/N_(DM). Here, ndm=1, . . . , N_(DM). F_(R)(D_(r), nud) canbe set in advance when Doppler aliasing range D_(r) and phase rotationamounts φ₁, φ₂, . . . , and φ_(N_DM) for applying Doppler shift amountsDOP₁, DOP₂, . . . , and DOP_(N_DM) are fixed. Therefore, for example,coded Doppler demultiplexer 212 may tabulate the correspondence between,on one hand, Doppler aliasing range D_(r) and the phase rotation amountsand, on the other hand, F_(R)(D_(r), nud), and may read F_(R)(D_(r),nud) based on Doppler aliasing range D_(r) and a phase rotation amount.Further, for example, when phase rotation amounts φ₁, φ₂, . . . , andφ_(N_DM) for applying Doppler shift amounts DOP₁, DOP₂, . . . , andDOP_(N_DM) satisfy −π≤φ₁<Φ₂< . . . <φ_(N_DM)<π, F_(R)(D_(r), nud) can beexpressed as in following Expression 45:

$\begin{matrix}{\mspace{79mu}\lbrack 48\rbrack} & \; \\{{F_{R}\left( {D_{r},{nud}} \right)} = {{{mod}\ \left( {{{nud} - 1 - D_{r}},N_{DM}} \right)} - {{ceil}\mspace{11mu}{\left( \frac{N_{DM}}{2} \right).}}}} & \left( {{Expression}\mspace{14mu} 45} \right)\end{matrix}$

For example, in accordance with Expressions 40 and 41, coded Dopplerdemultiplexer 212 calculates, within each range ofD_(r)∈{−ceil(Loc×N_(DM)/2), . . . , ceil(Loc×N_(DM)/2)−1}, receivedpower P_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(r), nuc, nud) after codeseparation using the unused coded Doppler multiplexed signal to whichDCI (nuc, nud) is assigned.

Then, coded Doppler demultiplexer 212 detects a D_(r) in which receivedpower P_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(r), nuc, nud) is minimizedamong the ranges of D_(r). In the following, D_(r) in which receivedpower P_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(r), nuc, nud) is minimizedamong the ranges of D_(r) as given by following Expression 46 isreferred to as “D_(r min)”:

$\begin{matrix}{\mspace{79mu}\lbrack 49\rbrack} & \; \\{D_{r_{\min}} = {\left\{ {{\arg\; D_{r}}❘{\min\limits_{D_{r}}{P_{DAR}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r},{nuc},{nud}} \right)}}} \right\}.}} & \left( {{Expression}\mspace{14mu} 46} \right)\end{matrix}$

Note that when there are a plurality of unused coded Doppler multiplexedsignals, coded Doppler demultiplexer 212 may use received powerPall_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(r)) after code separationusing all unused orthogonal codes as given by following Expression 47,instead of received power P_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(r),nuc, nud):

[50]

Pall _(DAR)(f _(b_cfar) ,f _(s_comp_cfar) ,D _(r))=Σ_(nuc,nud) P_(DAR)(f _(b_cfar) ,f _(s_comp_cfar) ,D _(r) ,nuc,nud)   (Expression47).

Obtaining the received power after code separation using all the unusedorthogonal codes makes it possible to increase the accuracy of thealiasing judgement by the aliasing processing even when the receptionsignal level is low.

For example, coded Doppler demultiplexer 212 calculatesPall_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(r)) in each range ofD_(r)∈{−ceil(Loc×N_(DM)/2), . . . , ceil(Loc×N_(DM)/2)−1}, and detectsD_(r) (in other words, D_(r) min) in which Pall_(DAR)(f_(b_cfar),f_(s_comp_cfar), D_(r)) is minimized. For example, when Expression 47 isused, D_(r) which provides the minimum received power among the rangesof D_(r) is represented as “D_(r min)” as given by following Expression48:

$\begin{matrix}{\mspace{79mu}\lbrack 51\rbrack} & \; \\{D_{r_{\min}} = {\left\{ {{\arg\; D_{r}}❘{\min\limits_{D_{r}}{{Pall}_{DAR}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r}} \right)}}} \right\}.}} & \left( {{Expression}\mspace{14mu} 48} \right)\end{matrix}$

Further, coded Doppler demultiplexer 212 may perform processing forjudging (in other words, measuring) the accuracy of the aliasingjudgement, for example, by comparing minimum received powerPall_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(r_min)) after code separationusing the unused coded Doppler multiplexed signal to which DCI (nuc,nud) is assigned, on the one hand, and received powerPowerFT_comp(f_(b_cfar), f_(s_comp_cfar)) of Expression 39 obtained inCFAR section 211 by performing power addition while adjusting peakpositions of Doppler-shift multiplexed signals, on the other hand. Inthis case, coded Doppler demultiplexer 212 may judge the accuracy of thealiasing judgement in accordance with following Expressions 49 and 50,for example:

[52]

Pall _(DAR)(f _(b_cfar) ,f _(s_comp_cfar) ,D_(r_min))<Threshold_(DR)×PowerFT_comp(f _(b_cfar) ,f _(s_comp_cfar))  (Expression 49);

[53]

Pall _(DAR)(f _(b_cfar) ,f _(s_comp_cfar) ,D_(r_min))≥Threshold_(DR)×PowerFT_comp(f _(b_cfar) ,f _(s_comp_cfar))  (Expression 50).

For example, when minimum received power Pall_(DAR)(f_(b_cfar),f_(s_comp_cfar), D_(rmin)) after code separation using the unused codedDoppler multiplexed signal to which DCI (nuc, nud) is assigned issmaller than the value obtained by multiplying, by predetermined valueThreshold_(DR), PowerFT_comp(f_(b), and f_(s_comp_cfar)) for distanceindex f_(b_cfar) and Doppler frequency index f_(s_comp_cfar) extractedin CFAR section 211 (for example, Expression 49), coded Dopplerdemultiplexer 212 judges that the aliasing judgement is sufficientlyaccurate. In this case, radar apparatus 10 may perform, for example,subsequent processing (e.g., code separation processing).

On the other hand, for example, when minimum received powerPall_(DAR)(f_(b_cfar), f_(s_comp_cfar), D_(rmin)) after code separationusing the unused coded Doppler multiplexed signal to which DCI (nuc,nud) is assigned is equal to or larger than the value obtained bymultiplying PowerFT_comp(f_(b), f_(s_comp_cfar)) by Threshold_(DR) (forexample, Expression 50), coded Doppler demultiplexer 212 judges that theaccuracy of the aliasing judgement is not sufficient and the reliabilityof the aliasing judgement is low (for example, noise component). In thiscase, for example, radar apparatus 10 may omit to perform subsequentprocessing (e.g., code separation processing).

Such processing makes it possible to reduce a judgement error inaliasing judgement and to remove a noise component. Note that,predetermined value Threshold_(DR) may, for example, be set to a rangeof from 0 to less than 1. By way of example, considering inclusion of anoise component, Threshold_(DR) may be set in a range of approximatelyfrom 0.1 to 0.5.

The operation example of the aliasing processing has been describedabove.

<(2) Doppler Code Separation Processing on Coded Doppler MultiplexedSignal Used for Multiplexing Transmission>

Coded Doppler demultiplexer 212 performs coded Doppler demultiplexingprocessing on a coded Doppler multiplexed signal used for multiplexingtransmission based on an aliasing judgement result.

For example, as given by following Expression 51, coded Dopplerdemultiplexer 212 applies Expression 41 based on D_(rmin) that is aresult of aliasing judgement in aliasing judgement processing, so as toseparate and receive the coded Doppler multiplexed signal to which DCI(ncm, ndm) used for multiplexing transmission is assigned. For example,coded Doppler demultiplexer 212 performs the separation processing usingfollowing Expression 51 to separate and receive the coded Dopplermultiplexed signal to which DCI (ncm, ndm) used for the multiplexingtransmission is assigned. Since by the aliasing judgement processing itis possible to judge an index (D_(rtrue)) that is a true Doppleraliasing range within the Doppler range of from −1/(2Tr) to less than1/(2Tr) (in other words, it is possible to judge an index such thatD_(rmin)=D_(rtrue)), it becomes possible for coded Doppler demultiplexer212 to set, to zero, the correlation value between the orthogonal codesused for code multiplexing in the Doppler range of from −1/(2Tr) to lessthan 1/(2Tr), so as to perform the separation processing in which theinterference between the code multiplexed signals is suppressed.

$\begin{matrix}{\mspace{85mu}\lbrack 54\rbrack} & \; \\{{Y_{z}\left( {f_{b\_ cfar},f_{{s\_ comp}{\_ cfar}},D_{rmin},{ncm},{ndm}} \right)} = {{Code}_{ncm}^{*}\left\{ {{\alpha\left( {f_{{s\_ comp}{\_ cfar}},D_{rnin}} \right)} \otimes {{VFTALL}_{z}\left( {f_{b\_ cfar},f_{{s\_ comp}{\_ cfar}},D_{rmin},{ndm}} \right)}} \right\}}} & \left( {{Expression}\mspace{14mu} 51} \right)\end{matrix}$

Here, Y_(z)(f_(b_cfar), f_(s_comp_cfar), D_(rmin), ncm, ndm) is anoutput (for example, coded Doppler demultiplexing result) resulting fromcode separation of the code multiplexed signal using orthogonal codeCode_(ncm) with respect to ndm-th coded Doppler multiplexed signalVFTALL_(z)(f_(b_cfar), f_(s_comp_cfar), D_(rmin), ndm) in Doppler rangeD_(rmin) among the outputs of distance indices f_(b_cfar) and Dopplerfrequency indices f_(s_comp_cfar) of Doppler analyzers 210 in z-thantenna system processor 201. It is possible to separate the codedDoppler multiplexed signal to which DCI (ncm, ndm) used for themultiplexing transmission is assigned. Note that, z=1, . . . , Na, andncm=1, . . . , N_(CM).

Through the code separation processing as described above, radarapparatus 10 can separate and receive the coded Doppler multiplexedsignal to which DCI (ncm, ndm) used for the multiplexing transmission isassigned. The separation and reception are based on the result of thealiasing judgement performed by Doppler analyzers 210 assuming a Dopplerrange of ±1/(2Tr) that is greater by a factor of Loc than the Dopplerrange of ±1/(2Loc×Tr) in which no aliasing occurs.

Further, since the coded Doppler multiplexed signal to which DCI (ncm,ndm) is assigned is transmitted from transmission antenna Tx #[ncm,ndm], it is also possible to judge transmission antenna 109. In otherwords, radar apparatus 10 can separate and receive the coded Dopplermultiplexed signal which is transmitted from transmission antenna Tx#[ncm, ndm] and to which DCI (ncm, ndm) is assigned.

In addition, for example, during the coded Doppler demultiplexingprocessing, radar apparatus 10 performs, on the outputs of Doppleranalyzer 210 for each code element, phase correction based on Dopplerphase correction taking into account Doppler aliasing (for example,Doppler phase correction vector α(f_(s_comp_cfar), D_(r))). Such phasecorrection corresponds to correcting phase changes corresponding toDoppler components among the Doppler component candidates with respectto f_(s_comp_cfar). Mutual interference between code multiplexed signalscan thus be reduced, for example, as low as a noise level. In otherwords, radar apparatus 10 can reduce inter-code interference to suppressan effect on degradation of the detection performance of radar apparatus10.

The foregoing description has been given of an example of the operationof coded Doppler demultiplexer 212.

In FIG. 1, peak extractor 213 outputs, to direction estimator 214, atleast one of the outputs of Doppler analyzers 210 for distance indexf_(b_cfar) and Doppler frequency index f_(s_comp_cfar) inputted fromCFAR section 211. At this time, peak extractor 213 may use, for example,D_(rmin) that is a Doppler aliasing judgement result inputted from codedDoppler demultiplexer 212.

For example, in the example illustrated in FIG. 1, peak extractor 213outputs output VFT_(z) ¹(f_(b_cfar),f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin), ndm__(BF))/N_(DM))) of firstDoppler analyzer 210 (Doppler analyzer 210-1) to direction estimator214. Here, ndm__(BF) is any one of 1, . . . , N_(DM), and a plurality oftransmission antennas 109 to which the ndm__(BF)-th Doppler multiplexedsignal is assigned are a combination of transmission antennas 109 thatsatisfies the condition of the adjacent arrangement described above, forexample.

In FIG. 1, based on Doppler aliasing judgement result D_(rmin) fordistance index f_(b_cfar) and Doppler frequency index f_(s_comp_cfar)inputted from coded Doppler demultiplexer 212, direction estimator 214performs direction estimation processing for estimation of the directionof a target based on separated received signal Y_(z)(f_(b_cfar),f_(s_comp_cfar), D_(rmin), ncm, ndm) of the coded Doppler multiplexedsignal to which DCI (ncm, ndm) is assigned and which is transmitted fromtransmission antenna Tx #[ncm, ndm], and based on the output from a partof Doppler analyzers 210 (Doppler analyzer 210-1 in FIG. 1) inputtedfrom peak extractor 213.

Note that, in the following description, a case in which output VFT_(z)¹(f_(b_cfar), f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin),ndm__(BF))/N_(DM))) from first Doppler analyzer 210 is used will bedescribed as an example, but the output from peak extractor 213 is notlimited to this. In addition, z=1, . . . , Na.

For example, direction estimator 214 generates, based on the outputs ofcoded Doppler demultiplexer 212 and peak extractor 213, virtualreception array correlation vector h(f_(b_cfar), f_(s_comp_cfar)) givenby following Expression 52 and performs the direction estimationprocessing.

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) includes Nt×Na elements, the number of which is theproduct of number Nt of transmission antennas and number Na of receptionantennas, and further includes an element resulting from use of a beamtransmission antenna. Detailed descriptions will be given below.

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) includes elements of beam transmission antennas. Theelements of beam transmission antennas are based on the output (e.g.,VFT_(z) ¹(f_(b_cfar), f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin),ndm__(BF))/N_(DM)))) of a part of Doppler analyzers 210 that is inputtedfrom peak extractor 213. The elements of beam transmission antennasresult from code multiplexing transmission using the same Dopplermultiplexing and constitute a sub-array by adjacent transmissionantennas 109 for orthogonal beam transmission.

For example, when there are N_(BF) beam transmission antennas, virtualreception array correlation vector h(f_(b_cfar), f_(s_comp_cfar))includes (Nt+N_(BF))×Na elements. By way of example, when number N_(BF)of beam transmission antennas is 1, virtual reception array correlationvector h(f_(b_cfar), f_(s_comp_cfar)) is expressed by followingExpression 52. Expression 52 represents an example in which peakextractor 213 outputs output VFT_(z) ¹(f_(b_cfar),f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin), ndm__(BF))/N_(DM))) from firstDoppler analyzer 210 to direction estimator 214, but the presentinvention is not limited to this.

Further, since the output of coded Doppler demultiplexer 212 and theoutput of peak extractor 213 have different noise levels, valuesobtained by multiplication by a normalizing factor may be used asvirtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)).

$\begin{matrix}\lbrack 55\rbrack & \; \\{{h\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}}} \right)} = \begin{Bmatrix}{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\{VF{T_{1}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{\begin{matrix}{N_{code}F_{R}} \\\left( {D_{r\min},{ndm\_ BF}} \right)\end{matrix}}{N_{DM}}}} \right)}} \\{{VF}{T_{2}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{\begin{matrix}{N_{code}F_{R}} \\\left( {D_{r\min},{ndm\_ BF}} \right)\end{matrix}}{N_{DM}}}} \right)}} \\\vdots \\{{VF}{T_{Na}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{\begin{matrix}{N_{code}F_{R}} \\\left( {D_{r\min},{ndm\_ BF}} \right)\end{matrix}}{N_{DM}}}} \right)}}\end{Bmatrix}} & \left( {{Expression}\mspace{14mu} 52} \right)\end{matrix}$

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) is used in processing for performing, on reflected wavesignals from a target, direction estimation based on a phase differencebetween reception antennas 202.

Note that, since the directivity pattern is different between the beamtransmission antenna and transmission antennas 109, it is preferable,for example, that direction estimator 214 perform the directionestimation processing in a range where the difference in directivitygain between the beam transmission antenna and transmission antennas 109is within a predetermined range.

[Antenna Arrangement Example]

For example, a description will be given of a case where number Nt oftransmission antennas used for multiplexing transmission is 3, numberN_(DM) of Doppler multiplexing is 2, number N_(CM) of code multiplexingis 2, and orthogonal code sequences Code₁={1, 1} and Code₂={1, −1} withcode length Loc=2 are set, and numbers N_(DOP_CODE)(1) andN_(DOP_CODE)(2) of coded Doppler multiplexing are set to 2 and 1,respectively. Note that number N_(BF) of beam transmission antennas isset to 1, and ndm__(BF)=1 is used as an index of a Doppler multiplexedsignal used for the beam transmission antenna.

In FIG. 17, for example, in radar apparatus 10, threehorizontally-arranged transmission antennas 109 (Tx #1, Tx #2, and Tx#3) are transmission antenna Tx #[1, 1], transmission antenna Tx #[2,1], and transmission antenna Tx #[1, 2] from the left. In FIG. 17, lefttwo adjacent transmission antennas Tx #1 (Tx #[1, 1]) and Tx #2 (Tx #[2,1]) transmit radar transmission signals using the same Dopplermultiplexing (Doppler shift amount=DOP₁). Thus, in FIG. 17, the beamtransmission antenna is formed by Tx #1 and Tx #2 (first sub-arrayantenna). In FIG. 17, number N_(BF) of beam transmission antennas is 1.In the following, the beam transmission antenna in FIG. 17 may also bereferred to as “Tx #4.”

Further, as illustrated in FIG. 17, number Na of reception antennas istwo (e.g., Rx #1 and Rx #2). Note that, number Na of reception antennasis not limited to two, and may be three or more, for example.

For example, when a radar transmission signal is transmitted fromadjacent Tx #1 (Tx #[1, 1]) and Tx #2 (Tx #[2, 1]), for example, at anequal power, the midpoint position between Tx #1 and Tx #2 serves as thephase center of beam transmission antenna Tx #4 (the cross markillustrated at (a) in FIG. 17). Note that, when the radar transmissionsignal is not transmitted at an equal power from transmission antennas109 constituting the beam transmission antenna, transmission at aposition dependent on the ratio of transmission powers of respectivetransmission antennas 109 constituting the beam transmission antenna(the position of the center of gravity of the transmission powers fromthe respective transmission antennas) that serves as the phase center ofthe sub-array can be treated as transmission by the beam transmissionantenna.

Arrangement of VA #1 to VA #8 of virtual reception antennas (or MIMOvirtual antennas) as illustrated at (b) in FIG. 17 is constituted by thearrangement of transmission antennas Tx #1 to Tx #3, beam transmissionantenna Tx #4 (e.g., Nt+N_(BF) transmission antennas), and receptionantennas Rx #1 and Rx #2 (e.g., Na reception antennas) as illustrated at(a) in FIG. 17. At (b) in FIG. 17, the virtual reception antennaarrangement obtained based on beam transmission antenna Tx #4corresponds to VA #7 and VA #8.

Here, the arrangement of the virtual reception antennas (the virtualreception array) may be expressed by following Expression 53, forexample, based on the positions of transmission antennas 109constituting the transmission array antenna (e.g., the positions offeeding points) and the positions of reception antennas 202 constitutingthe reception array antenna (e.g., the positions of feeding points):

$\begin{matrix}\lbrack 56\rbrack & \; \\\left\{ {\begin{matrix}{X_{{V\_}\# k} = {\left( {X_{{T\_}{\#{\lbrack{{ceil}{({k/{Na}})}}\rbrack}}} - X_{{T\_}{\# 1}}} \right) + \left( {X_{{R\_}{\#{\lbrack{{{mod}{({k - {1/{Na}}})}} + 1}\rbrack}}} - X_{{R\_}{\# 1}}} \right)}} \\{Y_{{V\_}\# k} = {\left( {Y_{{T\_}{\#{\lbrack{{ceil}{({k/{Na}})}}\rbrack}}} - Y_{{T\_}{\# 1}}} \right) + \left( {Y_{{R\_}{\#{\lbrack{{{mod}{({k - {1/{Na}}})}} + 1}\rbrack}}} - Y_{{R\_}{\# 1}}} \right)}}\end{matrix}.} \right. & \left( {{Expression}\mspace{14mu} 53} \right)\end{matrix}$

Here, the position coordinates of transmission antennas 109 (e.g., Tx#n) constituting the transmission array antenna are represented as (XT#11, YT #n) (e.g., n=1, . . . , Nt+N_(BF)), the position coordinates ofreception antennas 202 (e.g., Rx #m) constituting the reception arrayantenna are represented as (X_(R_#m), Y_(R_#m)) (e.g., m=1, . . . , Na),and the position coordinates of virtual antennas VA #k constituting avirtual reception array antenna are represented as (Xv #k, Y_(V_#k))(e.g., k=1, . . . , (Nt+N_(BF))×Na).

Note that, VA #1 is represented as the position reference (0, 0) of thevirtual reception array, for example, in Expression 53.

As illustrated at (b) in FIG. 17, the virtual reception antennaarrangement using the beam transmission antenna is the equally spacedarray arrangement of eight elements. On the other hand, as illustratedin FIG. 18, when no beam transmission antenna is used in the sameantenna arrangement as (a) in FIG. 17, number Nt of transmissionantennas is 3 and number Na of reception antennas is 2, and the virtualreception antenna arrangement is an equally spaced array arrangement ofsix elements when the equally spaced arrangement is formed in the samemanner as at (b) in FIG. 17.

Thus, the virtual reception antenna arrangement using the beamtransmission antenna makes it possible to enlarge the aperture length ofthe virtual reception antennas (e.g., increase the number of virtualreception antennas), so as to improve the angular resolution. Further,in the virtual reception antenna arrangement using the beam transmissionantenna, it is possible to suppress an increase in sidelobes by denselyarranging the virtual reception antennas to improve the angularresolution.

Although the example illustrated in FIG. 17 illustrates the case wherenumber Nt of transmission antennas is 3 and number N_(BF) of beamtransmission antennas is 1, number Nt of transmission antennas andnumber N_(BF) of beam transmission antennas are not limited thereto. Forexample, an increase in the number of transmission antennas 109 allowsfor the use of a larger number of beam transmission antennas, thusimproving the angular resolution or suppressing the sidelobe level. Notethat, the antennas illustrated in FIGS. 17 and 18 may be a part of aplurality of antennas that radar apparatus 10 includes.

The antenna arrangement example has been described above.

For example, direction estimator 214 calculates a spatial profile, withazimuth direction θ in direction estimation evaluation function valueP_(H) (θ, f_(b_cfar), f_(s_comp_cfar)) being variable within a definedangular range. Direction estimator 214 extracts a predetermined numberof local maximum peaks in the calculated spatial profile in descendingorder and outputs the azimuth directions of the local maximum peaks asdirection-of-arrival estimation values (for example, positioningoutputs).

Note that, there are various methods with direction estimationevaluation function value P_(H) (θ, f_(b_cfar), f_(s_comp_cfar))depending on direction-of-arrival estimation algorithms. For example, anestimation method using an array antenna, as disclosed in NPL 3, may beused.

For example, when (Nt+N_(BF))×Na virtual reception antennas are linearlyarranged at equal intervals d_(H), a beamformer method can be given byfollowing Expressions 54 and 55. In addition, a technique such as Caponor MUSIC is also applicable.

$\begin{matrix}{\mspace{79mu}\lbrack 57\rbrack} & \; \\{{P_{H}\left( {\theta_{u},f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}}} \right)} = {{{a^{H}\left( \theta_{u} \right)}D_{cal}{h\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}}} \right)}}}^{2}} & \left( {{Expression}\mspace{14mu} 54} \right) \\{\mspace{79mu}\lbrack 58\rbrack} & \; \\{{a\left( \theta_{u} \right)} = \begin{bmatrix}1 \\{\exp\left\{ {{- j}\; 2\;\pi\; d_{H}\sin\;\theta_{u}\text{/}\lambda} \right\}} \\\vdots \\{\exp\left\{ {{- j}\; 2\;\pi\;\left( {{\left( {N_{t} + N_{BF}} \right)N_{a}} - 1} \right)d_{H}\sin\;\theta_{u}\text{/}\lambda} \right\}}\end{bmatrix}} & \left( {{Expression}\mspace{14mu} 55} \right)\end{matrix}$

Here, in Expression 54, superscript H denotes the Hermitian transposeoperator.

Further, a(θ_(u)) denotes the direction vector of the virtual receptionarray relative to an incoming wave in azimuth direction θ_(u).

Azimuth direction θ_(u) is a vector that is changed at azimuth intervalpi in an azimuth range over which direction-of-arrival estimation isperformed. For example, θ_(u) is set as follows:

θ_(u)=θmin+uβ ₁ ,u=0, . . . ,NU;

NU=floor[(θmax−θmin)/β₁]+1.

Here, floor(x) is a function that returns the largest integer value notgreater than real number x.

Further, in Expression 54, Dcal is an ((Nt+N_(BF))×Na)-th order matrixincluding an array correction coefficient for correcting phasedeviations and amplitude deviations between the transmission arrayantennas and between the reception array antennas, and a coefficient forreducing the influence of coupling of elements between the antennas. Ifthe coupling between antennas in the virtual reception array isnegligible, Dcal is a diagonal matrix with diagonal components includingan array correction coefficient for correcting phase deviations andamplitude deviations between the transmission array antennas and betweenthe reception array antennas.

For example, direction estimator 214 may output, as a positioningresult, distance information based on distance index f_(b_cfar) andDoppler velocity information of the target based on the Dopplerfrequency judgement result for the target (result of Doppler aliasingjudgement processing by coded Doppler demultiplexer 212), together withthe direction estimation result. Direction estimator 214 may output thepositioning result, for example, to a vehicle control apparatus (notillustrated) when the radar apparatus is used as an in-vehicle radar, orto an infrastructure control apparatus (not illustrated) when the radarapparatus is used as an infrastructure radar.

Note that, for example, when the phase rotation amount is determinedusing Expression 5, the Doppler frequency information can be calculatedin the extended range as given by following Expression 56 using D_(rmin)that is the result of the Doppler aliasing judgement processing by codedDoppler demultiplexer 212:

$\begin{matrix}\lbrack 59\rbrack & \; \\{{f_{out} = {f_{{s\_ comp}{\_ cfar}} + \frac{D_{r\min}N_{code}}{N_{DM}}}}.} & \left( {{Expression}\mspace{14mu} 56} \right)\end{matrix}$

Further, for example, when the phase rotation amount is determined usingExpression 6, the Doppler frequency information can be calculated in theextended range as given by following Expression 57 using D_(rmin) thatis the result of the Doppler aliasing judgement processing:

$\begin{matrix}\lbrack 60\rbrack & \; \\{{f_{out} = {f_{{s\_ comp}{\_ cfar}} + \frac{D_{r\min}N_{code}}{N_{DM} + N_{int}}}}.} & \left( {{Expression}\mspace{14mu} 57} \right)\end{matrix}$

Note that, the Doppler frequency information may be converted into therelative velocity component and then outputted. Following Expression 58may be used to convert Doppler frequency index four to relative velocitycomponent v_(d)(f_(out)) using D_(rmin) that is the result of theDoppler aliasing judgement for a target. Here, λ is the wavelength ofthe carrier frequency of an RF signal outputted from a transmissionradio (not illustrated) (when a chirp signal is used, the wavelength atthe center frequency of the chirp signal is used). Further, Δf denotesthe Doppler frequency interval in FFT processing performed in Doppleranalyzer 210. For example, in the present embodiment,Δf=1/{N_(code)×Loc×T_(r)}.

$\begin{matrix}\lbrack 61\rbrack & \; \\{{v_{d}\left( f_{out} \right)} = {\frac{\lambda}{2}f_{out}\Delta_{f}}} & \left( {{Expression}\mspace{14mu} 58} \right)\end{matrix}$

As described above, in the present embodiment, radar apparatus 10applies phase rotation amounts corresponding to Doppler shift amountsand orthogonal code sequences to radar transmission signals to performmultiplexing transmission of the radar transmission signals (in otherwords, coded Doppler multiplexed signals) from a plurality oftransmission antennas 109. Further, at least one pair of adjacenttransmission antennas 109 transmit radar transmission signals to whichthe same Doppler multiplexing (Doppler shift amount) is applied, toperform code multiplexing transmission using the same Dopplermultiplexing.

It is thus possible to regard, as reception signals of orthogonal beamtransmission, reception signals for each transmission period thatcorrespond to the signals transmitted from the at least one pair ofadjacent transmission antennas 109. Accordingly, a beam transmissionantenna forming a sub-array is obtained by the at least one pair ofadjacent transmission antennas 109. For example, when radar transmissionsignals are transmitted from transmission antennas 109 constituting thebeam transmission antenna, for example, at equal power, the beamtransmission antenna can be treated as an antenna for which a midpointposition between transmission antennas 109 serves as the phase center ofthe sub-array. Note that, when the radar transmission signals are nottransmitted at an equal power from transmission antennas 109constituting the beam transmission antenna, transmission at a positiondependent on the ratio of transmission powers of respective transmissionantennas 109 constituting the beam transmission antenna (the position ofthe center of gravity of the transmission powers from the respectivetransmission antennas) that serves as the phase center of the sub-arraycan be treated as transmission by the beam transmission antenna.

Thus, according to the present embodiment, it is possible for radarapparatus 10 to utilize the transmission antennas such that the numberthereof is made greater than the number (Nt) of transmission antennas109 for multiplexing transmission. Further, for example, radar apparatus10 (radar receiver 200 (corresponding to the reception circuit))performs sensing processing of sensing a target object (target) usingvirtual reception antennas (e.g., (Nt+N_(BF))×Na virtual receptionantennas) constituted by a beam transmission antenna (e.g., N_(BF)antennas) formed by adjacent transmission antennas 109, a plurality of(e.g., Nt) transmission antennas 109, and a plurality of (e.g., Na)reception antennas 202. As is understood, radar apparatus 10 canincrease the number of virtual reception antennas by using one or morebeam transmission antennas and the plurality of transmission antennas109. It is thus possible to improve the angle resolution of directionestimator 214 of radar apparatus 10 or to reduce the sidelobe level.Such improvement of the angular measurement performance makes itpossible to improve target-object sensing accuracy of radar apparatus10.

Further, in the present embodiment, each of a plurality of transmissionantennas 109 is associated with a combination of the Doppler shiftamount (DOP_(ndm)) and the orthogonal code sequence (DOP_(ncm)) suchthat at least one of the Doppler shift amount (DOP_(ndm)) and theorthogonal code sequence (DOP_(ncm)) differs from combination tocombination. In the present embodiment, for example, encoder 107 may setthe numbers of multiplexing by the orthogonal code sequences (in otherwords, the numbers of codes) corresponding respectively to the Dopplershift amounts in the combinations of the Doppler shift amounts and theorthogonal code sequences such that the numbers of multiplexing aredifferent (in other words, such that the numbers of coded Dopplermultiplexing for respective Doppler multiplexed transmission signals arenot uniform), for example, using the equal-interval Doppler shift amountsetting including the maximum equal-interval Doppler shift amountsetting.

By setting the numbers of coded Doppler multiplexing for the respectiveDoppler multiplexed transmission signals non-uniformly, radar apparatus10 can judge, based on, for example, the received power of acode-separated signal for each coded Doppler multiplexed signal,transmission antenna 109 associated with the coded Doppler multiplexedsignal (in other words, the combination of the Doppler shift amount andthe orthogonal code sequence) and the presence or absence of Doppleraliasing. It is thus possible for radar apparatus 10 to appropriatelyjudge a Doppler frequency of a target even in the presence of Doppleraliasing.

Thus, according to the present embodiment, radar apparatus 10 can extendthe effective Doppler frequency bandwidth to 1/(Tr) (e.g., a Dopplerrange of ±1/(2Tr)) to extend the detection range for detecting a Dopplerfrequency (relative velocity) without ambiguity. Accordingly, radarapparatus 10 can improve the target-object sensing accuracy over a widerDoppler frequency range.

Further, in the present embodiment, encoder 107 may set the same numberof coded Doppler multiplexing for each Doppler multiplexed transmissionsignal (in other words, may set the numbers of coded Dopplermultiplexing for respective Doppler multiplexed signals uniformly), forexample, by using the equal-interval Doppler shift amount setting ofintervals narrower than the intervals of the maximum equal-intervalDoppler shift amount setting. By setting the numbers of coded Dopplermultiplexing for respective Doppler multiplexed transmission signalsuniformly, radar apparatus 10 can individually separate and receive thecoded Doppler multiplexed signals transmitted from a plurality oftransmission antennas 109 over a Doppler range of ±1/(2×Loc×Tr), forexample, by aliasing judgement processing in reception processing.

Further, in the present embodiment, encoder 107 may set the same numberof coded Doppler multiplexing for each Doppler multiplexed transmissionsignal (in other words, may set the numbers of coded Dopplermultiplexing for respective Doppler multiplexed signals uniformly), forexample, by using the maximum equal-interval Doppler shift amountsetting. By setting the numbers of coded Doppler multiplexing forrespective Doppler multiplexed transmission signals uniformly, radarapparatus 10 does not apply the aliasing judgement processing in thereception processing, for example. In addition, radar apparatus 10 canindividually separate and receive the coded Doppler multiplexed signalstransmitted from a plurality of transmission antennas 109 over a Dopplerrange of ±1/(2Loc×N_(DM)×Tr), for example.

Further, in the present embodiment, the coded Doppler multiplexing,which is performed using both Doppler multiplexing and coding, canreduce the number of Doppler multiplexing as compared with a case wherethe Doppler multiplexing is used without the coding in multiplexingtransmission. It is thus possible to increase the intervals of phaserotation amounts for applying Doppler shifts, so as to relieve theaccuracy requirements (phase modulation accuracy) for the phaseshifters, for example, and achieve the cost reduction effect of an RFsection, including reduction of the man-hours required for adjustment ofthe phase shifters.

Further, in the present embodiment, since coded Doppler multiplexing isperformed using both Doppler multiplexing and coding, radar apparatus 10performs, for each code element, Fourier frequency analysis (FFTprocessing) for Doppler frequency detection (relative velocitydetection). Accordingly, for example, in comparison to the Fourierfrequency analysis (FFT processing) for the Doppler frequency detection(relative velocity detection) using the Doppler multiplexing without thecoding in multiplexing transmission, the FFT size is (1/code length) andthe number of times of FFT processing is increased by (code length)times. For example, when the amount of FFT operation for FFT size Nc isroughly estimated to be Nc×log₂(Nc), the coded Doppler multiplexingaccording to the present embodiment has an operation amount ratio ofabout {Loc×Nc/Loc×log₂(Nc/Loc)}/{Nc×log₂(Nc)}=1−log₂(Loc)/log₂(Nc)relative to the FFT operation with Doppler multiplexing without thecoding. For example, in a case where Loc=2 and Nc=1024, the operationamount ratio is 0.9. The operation reduction effect on FFT processingcan be achieved, and the effect of simplification of the circuitconfiguration and cost reduction can also be achieved.

(Variation 1 of Embodiment 1)

Phase rotation amount φ_(ndm) for applying Doppler shift amountDOP_(ndm) is not limited to, for example, the value given by Expression5 and the like. For example, phase rotation amount φ_(ndm) may be avalue given by following Expression 59. Here, round(x) represents theround function that outputs a rounded integer value for real number x.Note that, the term “round(N_(code)/N_(DM))” is introduced so that thephase rotation amount is an integer multiple of the Doppler frequencyinterval in Doppler analyzer 210. In addition, in Expression 59, theangle is expressed in radian.

$\begin{matrix}\lbrack 62\rbrack & \; \\{\phi_{ndm} = {\frac{2\pi}{N_{code}}{round}\mspace{14mu}\left( \frac{N_{code}}{N_{DM}} \right)\left( {{ndm} - 1} \right)}} & \left( {{Expression}\mspace{14mu} 59} \right)\end{matrix}$

(Variation 2 of Embodiment 1)

Embodiment 1 has been described in relation to the case where Dopplershift setter 106 sets Doppler shift amount DOP_(ndm) for applying phaserotation amount φ_(ndm) (here, ndm=1, . . . , N_(DM)) assuming thatnumber N_(DM) of Doppler multiplexing is 2 or more, but the number ofDoppler multiplexing is not limited to this case and number N_(DM) ofDoppler multiplexing may also be set to 1.

In this case, for example, Doppler shift setter 106 sets Doppler shiftamount DOP₁ such that 0≤DOP₁<1/(Tr×Loc) is satisfied. Alternatively,Doppler shift setter 106 may set Doppler shift amount DOP₁, for example,such that −1/(2Tr×Loc)≤DOP₁<1/(2Tr×Loc) is satisfied.

Further, phase rotation amount φ₁ for applying Doppler shift amount DOP₁may also be assigned as in Expression 3.

Furthermore, the operation of encoder 107 in the case where numberN_(DM) of Doppler multiplexing is set to 1 is the same as the operationperformed in the case where number N_(DM) of Doppler multiplexing is setto 1 and N_(DOP_CODE)(1) is set to Nt in Embodiment 1. For example,encoder 107 uses orthogonal code sequences of number N_(CM)=Nt of codemultiplexing to set coded Doppler phase rotation amountψ_(ndop_code(1), 1)(m) given by Expression 10 for phase rotation amountφ₁ for applying Doppler shift amount DOP₁ inputted from Doppler shiftsetter 106, and outputs the coded Doppler phase rotation amount to phaserotator 108. Here, ndop_code(1)=1, . . . , Nt.

Subsequent operations of radar transmitter 100 are the same as those ofEmbodiment 1, and thus, the description thereof is omitted.

Note that, when number N_(DM) of Doppler multiplexing is set to 1, alltransmission antennas 109 transmit radar transmission signals using thesame Doppler multiplexing (Doppler shift amount DOP₁). Thus, thearrangement of transmission antennas 109 does not have to be associatedwith the assignment of the coded Doppler phase rotation amount. Inaddition, when number N_(DM) of Doppler multiplexing is set to 1, onebeam transmission antenna (N_(BF)=1) can be used. Accordingly, radarapparatus 10 can use (Nt+1)×Na virtual reception antennas with respectto number Nt of transmission antennas for multiplexing transmission.

Moreover, when number N_(DM) of Doppler multiplexing is set to 1, codedDoppler demultiplexer 212 in radar receiver 200 performs codedemultiplexing processing.

Hereinafter, a difference in the operation of radar receiver 200 ascompared with that in Embodiment 1 will be described.

CFAR section 211 does not apply the Doppler domain compression CFARprocessing when number N_(DM) of Doppler multiplexing is set to 1. Forexample, CFAR section 211 applies Expression 38 to adaptively set athreshold and outputs, to coded Doppler demultiplexer 212, distanceindex f_(b_cfar) and Doppler frequency index f_(s_cfar) that provide areceived power greater than the threshold, and received-powerinformation PowerFT(f_(b_cfar), f_(s_cfar)).

Coded Doppler demultiplexer 212 separates code-multiplexed transmissionsignals based on distance index f_(b_cfar) and Doppler frequency indexf_(s_cfar) inputted from CFAR section 211 and the outputs of Doppleranalyzers 210.

For example, when Doppler shift amount DOP₁ is 0, coded Dopplerdemultiplexer 212 separates and receives the code multiplexed signals asgiven by following Expression 60. For example, by performing thedemultiplexing processing based on Expression 60, coded Dopplerdemultiplexer 212 can separate and receive a transmission signal towhich Code_(ncm) used for code multiplexing transmission is assigned.

[63]

YC _(z)(f _(b_cfar) ,f _(s_cfar) ,ncm)=Code_(ncm)*{α(f_(s_cfar),0)⊗VFTTALLC _(z)(f _(b_cfar) ,f _(s_cfar))}   (Expression 60)

Here, YC_(z)(f_(b_cfar), f_(s_cfar), ncm) is an output resulting fromcode separation of the code multiplexed signal using orthogonal codeCode_(ncm) with respect to outputs VFTALLC_(z)(f_(b_cfar), f_(s_cfar))of Doppler analyzers 210 in z-th antenna system processor 201 fordistance index f_(b_cfar) and Doppler frequency index f_(s_comp_cfar).As is understood, coded Doppler demultiplexer 212 can separate andreceive the transmission signal to which Code_(ncm) is assigned. Notethat, z=1, . . . , Na and ncm=1, . . . , N_(CM).

For example, in the case of Doppler shift amount DOP₁≠0, coded Dopplerdemultiplexer 212 substitutes, for f_(s_cfar) of α(f_(b_cfar), 0) inExpression 60, the Doppler frequency index obtained by subtracting fromf_(s_cfar) the Doppler frequency index corresponding to Doppler shiftamount DOP₁ applied by radar transmitter 100. It is thus possible toseparate and receive the transmission signal to which Code_(ncm) usedfor the code multiplexing transmission is assigned, as in the case ofDoppler shift amount DOP₁=0.

It becomes possible for coded Doppler demultiplexer 212, for example, toperform the demultiplexing processing in a Doppler range of 1/(2×Loc×Tr)or more and less than 1/(2×Loc×Tr).

Note that, VFTTALLC_(z)(f_(b_cfar), f_(s_cfar)) in Expression 60 is arepresentation in vector format of an extracted component correspondingto distance index f_(b_cfar) and Doppler frequency index f_(s_cfar)extracted in CFAR section 211, the component being from among outputsVFT_(z) ^(noc)(f_(b), f_(s)) of Loc Doppler analyzers 210 in z-thantenna system processor 201, for example, as given by followingExpression 61. Here, noc=1, . . . , Loc.

[64]

VFTALLC _(z)(f _(b_cfar) ,f _(s_cfar))=[VFT _(z) ¹(f _(b_cfar) ,f_(s_cfar)) . . . VFT _(z) ^(L) ^(oc) (f _(b_cfar) ,f _(s_cfar))]^(T)  (Expression 61)

In addition, the phase correction by Doppler phase correction vector airα(f_(s_cfar), 0) corresponds to correction of phase changescorresponding to Doppler components of the Doppler component candidateswith respect to f_(s_cfar). Mutual interference between code multiplexedsignals can thus be reduced, for example, as low as a noise level. Inother words, radar apparatus 10 can reduce inter-code interference tosuppress the effect on degradation of the detection performance of radarapparatus 10.

The foregoing description has been given of an example of the operationof code demultiplexing processing of coded Doppler demultiplexer 212.

Peak extractor 213 outputs, to direction estimator 214, at least one ofthe outputs of Doppler analyzers 210 for distance index f_(b_cfar) andDoppler frequency index f_(s_cfar) inputted from CFAR section 211. Forexample, when the output of first Doppler analyzer 210 (for example,Doppler analyzer 210-1) is used, peak extractor 213 outputs VFT_(z)¹(f_(b_cfar), f_(s_cfar)) to direction estimator 214.

Direction estimator 214 performs the direction estimation processing forestimation of the direction of a target based on separated receivedsignal YC_(z)(f_(b_cfar), f_(s_cfar), ncm) of the code multiplexedsignal for distance index f_(b_cfar) and Doppler frequency indexf_(s_cfar) inputted from coded Doppler demultiplexer 212, and the outputof a part of Doppler analyzers 210 inputted from peak extractor 213.

Note that, by way of example, a case will be described below in whichoutput VFT_(z) ¹(f_(b_cfar), f_(s_cfar)) from first Doppler analyzer 210is used, but the output from peak extractor 213 is not limited to thisexample. In addition, z=1, . . . Na and ncm=1, . . . , N_(CM).

For example, direction estimator 214 generates, based on the outputsfrom coded Doppler demultiplexer 212 and peak extractor 213, virtualreception array correlation vector h(f_(b_cfar), f_(s_cfar)) given byfollowing Expression 62 and performs the direction estimationprocessing.

Virtual reception array correlation vector h(f_(b_cfar), f_(s_cfar))includes Nt×Na elements, the number of which is the product of number Ntof transmission antennas and number Na of reception antennas, andfurther includes elements resulting from use of beam transmissionantennas. Detailed descriptions will be given below.

Virtual reception array correlation vector h(f_(b_cfar), f_(s_cfar))includes elements of beam transmission antennas. The elements of beamtransmission antennas are based on the output (e.g., VFT_(z)¹(f_(b_cfar), f_(s_cfar)) of a part of Doppler analyzers 210 as inputtedfrom peak extractor 213. The elements of beam transmission antennasresult from code multiplexing transmission using the same Dopplermultiplexing and constitute a sub-array by adjacent transmissionantennas 109 for orthogonal beam transmission.

For example, since N_(BF)=1 for beam transmission antennas in the caseof N_(DM)=1, virtual reception array correlation vector h(f_(b_cfar),f_(s_cfar)) includes (Nt+1)×Na elements. By way of example, virtualreception array correlation vector h(f_(b_cfar), f_(s_cfar)) isexpressed by following Expression 62. Expression 62 expresses an examplein which peak extractor 213 outputs output VFT_(z) ¹(f_(b_cfar),f_(s_cfar)) from first Doppler analyzer 210 to direction estimator 214,but the present disclosure is not limited to this.

[65]

$\begin{matrix}{{h\left( {f_{{b\_}c{far}},f_{s\_ cfar}} \right)} = \begin{Bmatrix}{{YC}_{1}\left( {f_{{b\_}c{far}},f_{s\_ cfar},1} \right)} \\{{YC}_{2}\left( {f_{{b\_}c{far}},f_{s\_ cfar},1} \right)} \\\vdots \\{{YC}_{Na}\left( {f_{{b\_}c{far}},f_{s\_ cfar},1} \right)} \\{{YC}_{1}\left( {f_{{b\_}c{far}},f_{s\_ cfar},2} \right)} \\{{YC}_{2}\left( {f_{{b\_}c{far}},f_{s\_ cfar},2} \right)} \\\vdots \\{{YC}_{Na}\left( {f_{{b\_}c{far}},f_{s\_ cfar},2} \right)} \\\vdots \\{{YC}_{1}\left( {f_{{b\_}c{far}},f_{s\_ cfar},N_{CM}} \right)} \\{{YC}_{2}\left( {f_{{b\_}c{far}},f_{s\_ cfar},N_{CM}} \right)} \\\vdots \\{{YC}_{Na}\left( {f_{{b\_}c{far}},f_{s\_ far},N_{CM}} \right)} \\{VF{T_{1}^{1}\left( {f_{{b\_}c{far}},f_{s\_ cfar}} \right)}} \\{{VF}{T_{2}^{1}\left( {f_{{b\_}c{far}},f_{s\_ cfar}} \right)}} \\\vdots \\{{VF}{T_{Na}^{1}\left( {f_{{b\_}c{far}},f_{s\_ cfar}} \right)}}\end{Bmatrix}} & \left( {{Expression}\mspace{14mu} 62} \right)\end{matrix}$

Virtual reception array correlation vector h(f_(b_cfar), f_(s_cfar)) isused for processing of performing direction estimation on reflected wavesignals from a target based on a phase difference between receptionantennas 202.

Note that, since the directivity pattern is different between the beamtransmission antennas and transmission antennas 109, it is preferable,for example, that direction estimator 214 perform the directionestimation processing in a range where the difference in directivitygain between the beam transmission antennas and transmission antennas109 is within a predetermined range.

The direction estimation processing using virtual reception arraycorrelation vector h(f_(b_cfar), f_(s_cfar)) and subsequent operationsof direction estimator 214 are the same as those in Embodiment 1, andthus will not be described.

[Antenna Arrangement Example]

For example, a description will be given of a case where number Nt oftransmission antennas used for multiplexing transmission is 2, numberN_(DM) of Doppler multiplexing is 1, number N_(CM) of code multiplexingis 2, and orthogonal code sequences Code₁={1, 1} and Code₂={1, −1} withcode length Loc=2 are set, and number N_(DOP_CODE)(1) of coded Dopplermultiplexing is set to 2. Note that number N_(BF) of beam transmissionantennas is set to 1, and ndm__(BF)=1 is used as an index of a Dopplermultiplexed signal used for the beam transmission antenna.

In FIG. 19, for example, in radar apparatus 10, twohorizontally-arranged transmission antennas 109 (Tx #1 and Tx #2) aretransmission antenna Tx #[1, 1] and transmission antenna Tx #[2, 1] fromthe left. In FIG. 19, two adjacent transmission antennas Tx #1 (Tx #[1,1]) and Tx #2 (Tx #[2, 1]) (first sub-array antenna) transmit radartransmission signals using the same Doppler multiplexing (Doppler shiftamount=DOP₁). Accordingly, in FIG. 19, number N_(BF) of beamtransmission antennas is 1. In the following, the beam transmissionantenna in FIG. 19 may also be referred to as “Tx #3.” Note that atleast two transmission antennas included in the sub-array antenna neednot be at least two closest transmission antennas.

Further, as illustrated in FIG. 19, number Na of reception antennas istwo (e.g., Rx #1 and Rx #2). Note that, number Na of reception antennasis not limited to two, and may be three or more, for example.

For example, when a radar transmission signal is transmitted fromadjacent Tx #1 (Tx #[1, 1]) and Tx #2 (Tx #[2, 1]), for example, at anequal power, the midpoint position between Tx #1 and Tx #2 serves as thephase center of beam transmission antenna Tx #3 (the cross markillustrated at (a) in FIG. 19). Note that, when the radar transmissionsignal is not transmitted at an equal power from transmission antennas109 constituting the beam transmission antenna, transmission at aposition dependent on the ratio of transmission powers of respectivetransmission antennas 109 constituting the beam transmission antenna(the position of the center of gravity of the transmission powers fromthe respective transmission antennas) that serves as the phase center ofthe sub-array can be treated as transmission by the beam transmissionantenna.

Arrangement of VA #1 to VA #6 of virtual reception antennas (or MIMOvirtual antennas) as illustrated at (b) in FIG. 19 is constituted by thearrangement of transmission antennas Tx #1 and Tx #2, beam transmissionantennas Tx #3 (e.g., Nt+1 transmission antennas), and receptionantennas Rx #1 and Rx #2 (e.g., Na reception antennas) as illustrated at(a) in FIG. 19. At (b) in FIG. 19, the virtual reception antennaarrangement obtained based on beam transmission antenna Tx #3corresponds to VA #5 and VA #6.

As illustrated at (b) in FIG. 19, since (Nt+1)=3 and Na=2, the virtualreception antenna arrangement using the beam transmission antenna is theequally spaced array arrangement of six elements. On the other hand,when no beam transmission antenna is used in the same antennaarrangement as (a) in FIG. 19 and in a case (not illustrated) where theequally spaced arrangement is formed in the same manner as at (b) inFIG. 19, the virtual reception antenna arrangement is an equally spacedarray arrangement of four elements since number Nt of transmissionantennas is 2 and number Na of reception antennas is 2.

Thus, the virtual reception antenna arrangement using the beamtransmission antenna makes it possible to enlarge the aperture length ofthe virtual reception antennas (e.g., increase the number of virtualreception antennas), so as to improve the angular resolution. Further,in the virtual reception antenna arrangement using the beam transmissionantenna, it is possible to suppress an increase in sidelobes by denselyarranging the virtual reception antennas to improve the angularresolution.

Note that, the example illustrated in FIG. 19 illustrates the case wherenumber Nt of transmission antennas is 2, but number Nt of transmissionantennas is not limited thereto. For example, an increased number oftransmission antennas 109 allows use of a larger number of transmissionantennas 109 as beam transmission antennas, thereby increasing thedirectivity gain. For example, when number Nt of transmission antennasused for multiplexing transmission is 4, it is possible to use 4 (=Nt)transmission antennas 109 as the beam transmission antennas by settingnumber N_(DM) of Doppler multiplexing to 1 and number N_(CM) of codemultiplexing to 4 and by using the orthogonal code sequences with codelength Loc=4. Note that, the antennas illustrated in FIG. 19 may be apart of a plurality of antennas that radar apparatus 10 includes.

The antenna arrangement example has been described above.

As described above, radar apparatus 10 applies phase rotation amountscorresponding to orthogonal code sequences to radar transmission signalsto perform multiplexing transmission of radar transmission signals (inother words, code multiplexed signals) from a plurality of transmissionantennas 109. Further, since reception signals for each transmissionperiod can be regarded as reception signals of orthogonal beamtransmission, a beam transmission antenna forming a sub-array isobtained by a plurality of transmission antennas 109. For example, whenradar transmission signals are transmitted at equal power fromtransmission antennas 109 constituting the beam transmission antenna,the beam transmission antenna can be treated as a new antenna (beamtransmission antenna) for which a midpoint position between transmissionantennas 109 serves as the phase center of the sub-array. Note that,when the radar transmission signals are not transmitted at an equalpower from transmission antennas 109 constituting the beam transmissionantenna, transmission at a position dependent on the ratio oftransmission powers of respective transmission antennas 109 constitutingthe beam transmission antenna (the position of the center of gravity ofthe transmission powers from the respective transmission antennas) thatserves as the phase center of the sub-array can be treated astransmission by the beam transmission antenna.

Thus, it is possible for radar apparatus 10 to utilize the transmissionantennas such that the number thereof is made greater than the number(Nt) of transmission antennas 109 for multiplexing transmission.Further, for example, radar apparatus 10 (radar receiver 200) performssensing processing of sensing an target object (target) using virtualreception antennas (for example, (Nt+1)×Na virtual reception antennas)that are constituted by a beam transmission antenna formed by aplurality of transmission antennas 109, a plurality of transmissionantennas 109, and a plurality of reception antennas 202. As isunderstood, it is possible for radar apparatus 10, for example, toincrease the virtual reception antennas by using the beam transmissionantenna and a plurality of transmission antennas 109 (e.g., it ispossible to utilize (Nt+1)×Na virtual reception antennas with respect tonumber Nt of transmission antennas for multiplexing transmission). It isthus possible to improve the angular resolution in direction estimator214 of radar apparatus 10 or reduce the sidelobe level. Such improvementof the angular measurement performance makes it possible to improvetarget-object sensing accuracy of radar apparatus 10.

(Variation 3 of Embodiment 1)

Variation 3 of Embodiment 1 will be described in relation to processing(for example, reception processing) in which encoder 107 sets thenumbers of coded Doppler multiplexing for Doppler multiplexed signalsuniformly by using the equal-interval Doppler shift amount setting ofintervals narrower than the intervals of the maximum equal-intervalDoppler shift amount setting.

For example, the following description will be given of an operation ofradar receiver 200 in a case where Doppler shift setter 106 sets thenumbers of coded Doppler multiplexing for the Doppler multiplexedsignals uniformly by using the equal-interval Doppler shift amountsetting of intervals narrower than the intervals of the maximumequal-interval Doppler shift amount setting (e.g., Expression 6).Hereinafter, a difference in the operation of radar receiver 200 betweenthe present embodiment and Embodiment 1 will be described.

For example, when the equal-interval Doppler shift amount setting usingphase rotation amount φ_(ndm) given by Expression 6 is used, N_(DM)peaks are detected at intervals of ΔFD=Ncode/(N_(DM)+N_(int)). It isthus possible for CFAR section 211 to apply the Doppler domaincompression CFAR processing. For example, as given by followingExpression 63, CFAR section 211 performs power addition for Dopplermultiplexed signals while adjusting peak positions of the Dopplermultiplexed signals to perform the Doppler domain compression CFARprocessing. Here, f_(s_comp)=ΔFD/2, . . . ,ΔFD/2−1=Ncode/{2(N_(DM)+N_(int))}, . . . , Ncode/{2(N_(DM)+N_(int))}−1.

$\begin{matrix}{\mspace{79mu}\lbrack 66\rbrack} & \; \\{{{Powe}rF{T_{comp}\left( {f_{b},f_{s_{comp}}} \right)}} = {\sum\limits_{{nfd} = 1}^{N_{DM} + N_{int}}\;{{PowerFT}\left( {f_{b},{f_{s_{-}comp}\  + {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right) \times \Delta{FD}}}} \right)}}} & \left( {{Expression}\mspace{14mu} 63} \right)\end{matrix}$

Note that, in Expression 63, in a case where

$\begin{matrix}\lbrack 67\rbrack & \; \\{{{f_{s_{-}comp}\  + {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right) \times \Delta\;{FD}}} < {{- {Ncode}}\text{/}2}},} & \;\end{matrix}$

the Doppler frequency index to which Ncode is added is used.

In addition, in Expression 63), in a case where

$\begin{matrix}\lbrack 68\rbrack & \; \\{{{f_{s_{-}comp}\  + {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right) \times \Delta\;{FD}}} > {\frac{Ncode}{2} - 1}},} & \;\end{matrix}$

the Doppler frequency index from which Ncode is further subtracted isused.

It is thus possible to compress the Doppler frequency range for the CFARprocessing to 1/(N_(DM)+N_(int)) to reduce the amount of CFAR processingand to simplify the circuit configuration. In addition, CFAR section 211is capable of power addition for N_(DM) Doppler-shift multiplexedsignals, to improve SNR by about (N_(DM))^(1/2). As a result, the radarsensing performance of radar apparatus 10 can be improved.

CFAR section 211 using the Doppler domain compression CFAR processingadaptively sets a threshold, for example, and outputs, to coded Dopplerdemultiplexer 212, distance index f_(b_cfar) and Doppler frequency indexf_(s_comp_cfar) that provide a received power greater than thethreshold, and received power information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD (where nfd=1, . . ., N_(DM)+N_(int))) for the Doppler frequency indices(f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) of N_(DM) Dopplermultiplexed signals.

Next, an example of the operation of coded Doppler demultiplexer 212illustrated in FIG. 1 will be described. The following describes anexample of processing performed by coded Doppler demultiplexer 212 whenCFAR section 211 uses the Doppler domain compression CFAR processing.

Based on the outputs of CFAR section 211 (e.g., distance indicesf_(b_cfar), Doppler frequency indices f_(s_comp_cfar), and receivedpower information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD (where nfd=1, . . ., (N_(DM)+N_(int)))) for the Doppler frequency indices(f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) of(N_(DM)+N_(int)) Doppler multiplexed signals), coded Dopplerdemultiplexer 212 separates the coded Doppler multiplexed transmissionsignals using the outputs of Doppler analyzers 210, and distinguishes(in other words, judges or identifies) transmission antennas 109 and theDoppler frequencies (in other words, the Doppler velocities or relativevelocities).

As described above, in the case where the numbers of coded Dopplermultiplexing for the Doppler multiplexed signals are set uniformly byusing the equal-interval Doppler shift amount setting of intervalsnarrower than the intervals of the maximum equal-interval Doppler shiftamount setting (e.g., Expression 6), coded Doppler demultiplexer 212performs, for example, (1) the aliasing judgement, and (2) the Dopplercode separation processing on the coded Doppler multiplexed signals usedfor multiplexing transmission based on the result of the aliasingjudgement.

Processing (1) and processing (2) by coded Doppler demultiplexer 212described above will be described below.

<(1) Aliasing Judgement Processing (Detection Processing of DetectingUnused Coded Doppler Multiplexed Signal)>

For example, in the aliasing judgement, coded Doppler demultiplexer 212detects N_(DM) peaks at intervals of ΔFD=Ncode/(N_(DM)+N_(int)) usingthe outputs of CFAR section 211 (e.g., distance indices f_(b_cfar),Doppler frequency indices f_(s_comp_cfar), and received powerinformation PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD (where nfd=1, . . ., (N_(DM)+N_(int)))) for the Doppler frequency indices(f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD of(N_(DM)+N_(int)) Doppler multiplexed signals). For example, usingreceived power information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) for Dopplerfrequency indices (f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD)of the (N_(DM)+N_(int)) Doppler multiplexed signals, coded Dopplerdemultiplexer 212 detects the Doppler frequency indices of N_(int) codedDoppler multiplexed signals not used for multiplexing transmission.Through this processing, coded Doppler demultiplexer 212 performs thealiasing judgement in the Doppler range of ±1/(2Loc×Tr).

For example, when Expression 6 is used and when N_(int)=1 is set in thecase of N_(DM)=3,

$\begin{matrix}\lbrack 69\rbrack & \; \\{\phi_{ndm} = {\frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM} + N_{int}} = \frac{2{\pi\left( {{ndm} - 1} \right)}}{4}}} & \;\end{matrix}$

holds true, and φ₁, φ₂, and φ₃(=(φ_(N_DM)) are 0°, 90°, and 180°,respectively. Here, ndm=1, . . . , N_(DM).

Further, the Doppler shift amounts corresponding respectively to suchphase rotations are DOP₁=0, DOP₂=ΔFD, and DOP₃ (=DOP_(N_DM))=2ΔFD,respectively. Thus, N_(DM) Doppler multiplexed signals are assigned atintervals of ΔFD, and there are N_(int) Doppler shift amounts that arenot assigned at the intervals of ΔFD. Here, N_(int)=1, and no Dopplermultiplexed signal is assigned to a Doppler shift amount of ΔFD.Further, radar apparatus 10 can identify N_(DM) Doppler multiplexedsignals (DOP₁, DOP₂, and DOP₃ (=DOP_(N_DM)) assigned at the intervals ofΔFD when successfully detecting the Doppler shift amounts that are notassigned at the intervals of ΔFD.

Signals holding the relationship of the Doppler shift amounts applied inradar transmitter 100 as described above are received as radar receptionsignals in radar apparatus 10. As is understood from the above, by usingthe outputs of CFAR section 211 (e.g., distance indices f_(b_cfar),Doppler frequency indices f_(s_comp_cfar), and received powerinformation PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD (where nfd=1, . . ., (N_(DM)+N_(int)))) for the Doppler frequency indices(f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD of(N_(DM)+N_(int)) Doppler multiplexed signals), coded Dopplerdemultiplexer 212 detects the Doppler frequency indices of N_(int) codedDoppler multiplexed signals not used for Doppler multiplexingtransmission. It is thus possible for coded Doppler demultiplexer 212 toperform the aliasing judgement in the Doppler range of ±1/(2Loc×Tr).

Here, the detection of the Doppler frequency indices of N_(int) codedDoppler multiplexed signals not used for Doppler multiplexingtransmission may be performed using received power informationPowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) as follows.

For example, when N_(int)=1, coded Doppler demultiplexer 212 detects aD_(r) in which received power PowerFT(f_(b_cfar),f_(s_comp_cfar)+(D_(r)−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) is minimizedamong the D_(r) ranges, as given by following Expression 64. Such aD_(r) is expressed as “D_(r min)” Here, D_(r) is an integer value in arange of D_(r)=−ceil((N_(DM)+N_(int))/2), ceil((N_(DM)+N_(int))/2)−1.

$\begin{matrix}{\mspace{79mu}\lbrack 70\rbrack} & \; \\{D_{r_{\min}} = \left\{ {{\arg\; D_{r}}❘{\min\limits_{D_{r}}{{PowerFT}\left( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\left( {D_{r} - {{ceil}\left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right)\Delta\;{FD}}}} \right)}}} \right\}} & \left( {{Expression}\mspace{14mu} 64} \right)\end{matrix}$

For example, when N_(int)>2, coded Doppler demultiplexer 212 detectsD_(r) of minimized power by utilizing the beforehand knowledge ofrelative positional relationship between the Doppler frequency indicesof N_(int) coded Doppler multiplexed signals not used for Dopplermultiplexing transmission in respective D_(r). For example, whenN_(int)>2, coded Doppler demultiplexer 212 detects a D_(r) in which thereceived power is minimized among the D_(r) using following Expression65. Such a D_(r) is expressed as “D_(r min).” Here, D_(r) is an integervalue in a range of D_(r)=−ceil((N_(DM)+N_(int))/2), . . . ,ceil((N_(DM)+N_(int))/2)−1. Here, F_(nint)(D_(r)) is an indexrepresenting the relative positional relationship of the Dopplerfrequency index of the nint-th coded Doppler multiplexed signal not usedfor Doppler multiplexing transmission in D_(r). Note that the indexinterval between that indices represented by F_(nint)(D_(r)) is ΔFD.Here, nint=1, . . . , N_(int).

$\begin{matrix}{\mspace{79mu}\lbrack 71\rbrack} & \; \\{D_{r\min} = \left\{ {{\arg\; D_{r}}❘{\min\limits_{D_{r}}{\sum\limits_{{nint} = 1}^{N_{int}}{{PowerFT}\left( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \ {\left( {{F_{nint}\left( D_{r} \right)} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right)\Delta FD}}} \right)}}}} \right\}} & \left( {{Expression}\mspace{14mu} 65} \right)\end{matrix}$

Coded Doppler demultiplexer 212 outputs, to peak extractor 213, analiasing judgement result (e.g., f_(b_cfar), f_(s_comp_cfar), D_(rmin))with respect to a reception signal for f_(b_cfar) and f_(s_comp_cfar).

The operation example of the aliasing processing has been describedabove.

<(2) Doppler Code Separation Processing on Coded Doppler MultiplexedSignal Used for Multiplexing Transmission>

Coded Doppler demultiplexer 212 performs coded Doppler demultiplexingprocessing on a coded Doppler multiplexed signal used for multiplexingtransmission based on an aliasing judgement result.

For example, based on Expression 51 and based on D_(rmin) which is aresult of aliasing judgement in aliasing judgement processing, the codedDoppler demultiplexer separates and receives the coded Dopplermultiplexed signal to which DCI (ncm, ndm) used for multiplexingtransmission is assigned. For example, coded Doppler demultiplexer 212can perform the separation processing using Expression 51 to separateand receive the coded Doppler multiplexed signal to which DCI (ncm, ndm)used for the multiplexing transmission is assigned.

Note that, following Expression 66 is used for VFTALL_(z)(f_(b_cfar),f_(s_comp_cfar), D_(r), ndm) in Expression 51. Note that, the term“N_(code)F_(R)(D_(r), ndm)/(N_(DM)+N_(int))” in Expression 66 can alsobe expressed as F_(R)(D_(r), ndm)ΔFD using ΔFD=Ncode/(N_(DM)+N_(int)).Therefore, the expression is applicable in the other cases than the caseof ΔFD=Ncode/(N_(DM)+N_(int)). In addition, the term“Ncode/(N_(DM)+N_(int))” in the following expressions represents ΔFD.When ΔFD=Ncode/(N_(DM)+N_(int)) is not used, the expression isapplicable and the same effect is achievable by replacingNcode/(N_(DM)+N_(int)) with ΔFD. Here, ndm=1, . . . , N_(DM).

$\begin{matrix}{\mspace{79mu}\lbrack 72\rbrack} & \; \\{{{VFTA}L{L_{z}\left( {f_{b\_ cfar},f_{{s\_ comp}{\_ cfar}},D_{r},{ndm}} \right)}} = {\quad{\quad\left\lbrack {{{VFT}_{z}^{1}\left( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \frac{\begin{matrix}{N_{code}F_{R}} \\\left( {D_{r},{ndm}} \right)\end{matrix}}{N_{DM} + N_{int}}}} \right)}\mspace{14mu}\left. \quad{\ldots\mspace{14mu}{{VFT}_{z}^{L_{oc}}\left( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \frac{\begin{matrix}{N_{code}F_{R}} \\\left( {D_{r},{ndm}} \right)\end{matrix}}{N_{DM} + N_{int}}}} \right)}} \right\rbrack^{T}} \right.}}} & \left( {{Expression}\mspace{14mu} 66} \right)\end{matrix}$

In Expression 66, F_(R)(D_(r), ndm) can be set in advance when Doppleraliasing range D_(r) and phase rotation amounts φ₁, φ₂, . . . , andφ_(N_DM) for applying Doppler shift amounts DOP₁, DOP₂, . . . , andDOP_(N_DM) are fixed. Therefore, for example, coded Dopplerdemultiplexer 212 may tabulate the correspondence between, on one hand,Doppler aliasing range D_(r) and the phase rotation amounts and, on theother hand, F_(R)(D_(r), ndm) and read F_(R)(D_(r), ndm) based onDoppler aliasing range D_(r) and a phase rotation amount. Further, forexample, when phase rotation amounts φ₁, φ₂, . . . , and φ_(N_DM) forapplying Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM)satisfy −π≤φ₁<φ₂< . . . <φ_(N_DM)<π, F_(R)(D_(r), ndm) can be expressedas in following

Expression 67:

$\begin{matrix}{\mspace{79mu}\lbrack 73\rbrack} & \; \\{{F_{R}\left( {D_{r},{ndm}} \right)} = {{{mod}\ \left( {{{ndm} - 1 - D_{r}},N_{DM}} \right)} - {{ceil}\mspace{11mu}{\left( \frac{N_{DM} + N_{int}}{2} \right).}}}} & \left( {{Expression}\mspace{14mu} 67} \right)\end{matrix}$

Since by the aliasing judgement processing it is possible to judge anindex (D_(rtrue)) that is a true Doppler aliasing range within theDoppler range of from −1/(2Loc×Tr) to less than 1/(2Loc×Tr) (in otherwords, it is possible to judge an index such that D_(rmin)=D_(rtrue)),it becomes possible for coded Doppler demultiplexer 212 to set, to zero,the correlation value between the orthogonal codes used for codemultiplexing in the Doppler range of from −1/(2Loc×Tr) to less than1/(2Loc×Tr), so as to perform the separation processing in which theinterference between the code multiplexed signals is suppressed.

Through the code separation processing as described above, and, based onthe aliasing judgement result assuming the Doppler range of±1/(2Loc×Tr), radar apparatus 10 can separate and receive the codedDoppler multiplexed signal to which DCI (ncm, ndm) used for themultiplexing transmission is assigned.

Further, since the coded Doppler multiplexed signal to which DCI (ncm,ndm) is assigned is transmitted from transmission antenna Tx #[ncm,ndm], it is also possible to judge transmission antenna 109. In otherwords, radar apparatus 10 can separate and receive the coded Dopplermultiplexed signal which is transmitted from transmission antenna Tx#[ncm, ndm] and to which DCI (ncm, ndm) is assigned.

In addition, for example, during coded Doppler demultiplexingprocessing, radar apparatus 10 performs, on the outputs of Doppleranalyzers 210 for each code element, Doppler phase correction, forexample, based on Doppler phase correction vector α(f_(s_comp_cfar),D_(r)) taking into consideration Doppler aliasing. Such phase correctioncorresponds to correcting phase changes corresponding to Dopplercomponents among the Doppler component candidates with respect tof_(s_comp_cfar). Mutual interference between code multiplexed signalscan thus be reduced, for example, as low as a noise level. In otherwords, radar apparatus 10 can reduce inter-code interference to suppressthe effect on degradation of the detection performance of radarapparatus 10.

The foregoing description has been given of an example of the operationof coded Doppler demultiplexer 212.

In FIG. 1, peak extractor 213 outputs, to direction estimator 214, atleast one of the outputs of Doppler analyzers 210 for distance indexf_(b_cfar) and Doppler frequency index f_(s_comp_cfar) inputted fromCFAR section 211. At this time, peak extractor 213 may use, for example,D_(rmin) that is a Doppler aliasing judgement result inputted from codedDoppler demultiplexer 212.

For example, in the example illustrated in FIG. 1, peak extractor 213outputs output VFT_(z) ¹(f_(b_cfar),f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin), ndm__(BF))/(N_(DM)+N_(int))))of first Doppler analyzer 210 (Doppler analyzer 210-1)) to directionestimator 214. Here, ndm__(BF) is any one of 1, . . . , N_(DM), and aplurality of transmission antennas 109 to which the ndm__(BF)-th Dopplermultiplexed signal is assigned are a combination of transmissionantennas 109 that satisfies the condition of the adjacent arrangementdescribed above, for example.

In FIG. 1, based on aliasing judgement result D_(rmin) for distanceindex f_(b_cfar) and Doppler frequency index f_(s_comp_cfar) inputtedfrom coded Doppler demultiplexer 212, direction estimator 214 performsdirection estimation processing for estimation of the direction of atarget based on separated received signal Y_(z)(f_(b_cfar),f_(s_comp_cfar), D_(rmin), ncm, ndm) of the coded Doppler multiplexedsignal to which DCI (ncm, ndm) is assigned and which is transmitted fromtransmission antenna Tx #[ncm, ndm], and based on the output from a partof Doppler analyzers 210 (Doppler analyzer 210-1 in FIG. 1) inputtedfrom peak extractor 213.

Note that, by way of example, the case where output VFT_(z)¹(f_(b_cfar), f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin),ndm__(BF))/(N_(DM)+N_(int)))) from first Doppler analyzer 210 is usedwill be described below, but the output from peak extractor 213 is notlimited to this. In addition, z=1, . . . , Na.

For example, direction estimator 214 generates, based on the outputs ofcoded Doppler demultiplexer 212 and peak extractor 213, virtualreception array correlation vector h(f_(b_cfar), f_(s_comp_cfar)) givenby following Expression 68 and performs the direction estimationprocessing.

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) includes Nt×Na elements, the number of which is theproduct of number Nt of transmission antennas and number Na of receptionantennas, and further includes elements resulting from use of beamtransmission antennas. Detailed descriptions will be given below.

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) includes elements of beam transmission antennas. Theelements of beam transmission antennas are based on the output (e.g.,VFT_(z) ¹(f_(b_cfar), f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin),ndm__(BF))/(N_(DM)+N_(int))))) of a part of Doppler analyzers 210 thatis inputted from peak extractor 213. The elements of beam transmissionantennas result from code multiplexing transmission using the sameDoppler multiplexing and constitute a sub-array by adjacent transmissionantennas 109 for orthogonal beam transmission.

For example, when there are N_(BF) beam transmission antennas, virtualreception array correlation vector h(f_(b_cfar), f_(s_comp_cfar))includes (Nt+N_(BF))×Na elements. By way of example, when number N_(BF)of beam transmission antennas is 1, virtual reception array correlationvector h(f_(b_cfar), f_(s_comp_cfar)) is expressed by followingExpression 68. In Expression 68, an example is expressed in which peakextractor 213 outputs output VFT_(z) ¹(f_(b_cfar),f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin), ndm__(BF))/N_(DM)+N_(int))))from first Doppler analyzer 210 to direction estimator 214, but thepresent invention is not limited to this.

Further, since the output of coded Doppler demultiplexer 212 and theoutput of peak extractor 213 have different noise levels, valuesobtained by multiplication by a normalizing factor may be used asvirtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)).

$\begin{matrix}\lbrack 74\rbrack & \; \\{{h\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}}} \right)} = \begin{Bmatrix}{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\{VF{T_{1}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{ndm\_ BF}} \right)}}{N_{DM} + N_{int}}}} \right)}} \\{{VF}{T_{2}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{ndm\_ BF}} \right)}}{N_{DM} + N_{int}}}} \right)}} \\\vdots \\{{VF}{T_{Na}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{ndm\_ BF}} \right)}}{N_{DM} + N_{int}}}} \right)}}\end{Bmatrix}} & \left( {{Expression}\mspace{14mu} 68} \right)\end{matrix}$

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) is used in processing for performing, on reflected wavesignals from a target, direction estimation based on a phase differencebetween reception antennas 202.

Since subsequent operations of direction estimator 214 are the same asthose in Embodiment 1, the description thereof is omitted.

The foregoing description has been given above of the operation exampleof radar receiver 200 in the case where encoder 107 sets the numbers ofcoded Doppler multiplexing for Doppler multiplexed signals uniformly byusing the equal-interval Doppler shift amount setting of intervalsnarrower than the intervals of the maximum equal-interval Doppler shiftamount setting.

[Antenna Arrangement Example]

Hereinafter, an example of arrangement of antennas in the case ofuniformly setting the numbers of coded Doppler multiplexing. Further,for example, an example of an antenna arrangement in the case where thenumbers of coded Doppler multiplexing for Doppler multiplexed signalsare uniformly set and a plurality of numbers of beam transmissionantennas are set.

For example, with reference to FIG. 20, a description will be given of acase where number Nt of transmission antennas used for multiplexingtransmission is 4, number N_(DM) of Doppler multiplexing is 2, N_(CM) is2, orthogonal code sequences Code₁ {1, 1} and Code₂ {1, −1} with codelength Loc=2 are set, and numbers N_(DOP_CODE)(1) and N_(DOP_CODE)(2) ofcoded Doppler multiplexing are 2 and 2, respectively, in radar apparatus10. Note that number N_(BF) of beam transmission antennas is set to 2,and ndm__(BF1)=1 and ndm__(BF2)=2 are used as indices of Dopplermultiplexed signals used for the beam transmission antennas.

In FIG. 20, for example, horizontally arranged four transmissionantennas 109 (Tx #1, Tx #2, Tx #3, and Tx #4) are transmission antennaTx #[1, 1], transmission antenna Tx #[2, 1], transmission antenna Tx#[1, 2], and transmission antenna Tx #[2, 2] from the left. In FIG. 20,two transmission antennas Tx #1 (Tx #[1, 1]) and Tx #2 (Tx #[2, 1])(first sub-array antenna) transmit radar transmission signals using thesame Doppler multiplexing (Doppler shift amount=DOP₁). Further, twoadjacent transmission antennas Tx #3 (Tx #[1, 2]) and Tx #4 (Tx #[2, 2])(second sub-array antenna) transmit radar transmission signals using thesame Doppler multiplexing (Doppler shift amount=DOP₂). Thus, in FIG. 20,one beam transmission antenna is formed by Tx #1 and Tx #2, and one beamtransmission antenna is formed by Tx #3 and Tx #4. In FIG. 20, numberN_(BF) of beam transmission antennas is 2. In the following, the beamtransmission antenna corresponding to Tx #1 and Tx #2 may also bereferred to as “Tx #5,” and the beam transmission antenna correspondingto Tx #3 and Tx #4 may also be referred to as “Tx #6” in FIG. 20.

Further, in FIG. 20, number Na of reception antennas is two (e.g., Rx #1and Rx #2). Note that, number Na of reception antennas is not limited totwo, and may be three or more, for example.

For example, when radar transmission signals are transmitted fromadjacent Tx #1 (Tx #[1, 1]) and Tx #2 (Tx #[2, 1]) and from adjacent Tx#3 (Tx #[1, 2]) and Tx #4 (Tx #[2, 2]), for example, at an equal power,the midpoint position between Tx #1 and Tx #2 serves as the phase centerof beam transmission antenna Tx #5, and the midpoint position between Tx#3 and Tx #4 serve as the phase center of beam transmission antenna Tx#6 (the cross marks illustrated at (a) in FIG. 20). Note that, when theradar transmission signals are not transmitted at an equal power fromtransmission antennas 109 constituting the beam transmission antennas,transmission at each position that is dependent on the ratio oftransmission powers of respective transmission antennas 109 constitutingthe beam transmission antenna (the position of the center of gravity ofthe transmission powers from the respective transmission antennas) andserves as the phase center of the sub-array can be treated astransmission by the beam transmission antenna.

Arrangement of VA #1 to VA #12 of virtual reception antennas (or MIMOvirtual antennas) as illustrated at (b) in FIG. 20 is constituted by thearrangement of transmission antennas Tx #1 to Tx #4, beam transmissionantennas Tx #5 and Tx #6, and reception antennas Rx #1 and Rx #2 asillustrated at (a) in FIG. 20. At (b) in FIG. 20, the virtual antennaarrangement obtained based on beam transmission antennas Tx #5 and Tx #6corresponds to VA #9, VA #11, VA #10, and VA #12.

Here, the arrangement of the virtual reception array may be expressed byfollowing Expression 53, for example, based on the positions oftransmission antennas 109 constituting the transmission array antenna(e.g., the positions of the feeding points) and the position ofreception antenna 202 constituting the reception array antenna (e.g.,the position of the feeding point).

As illustrated at (b) in FIG. 20, since (Nt+N_(BF))=6 and Na=2, thevirtual reception antenna arrangement using the beam transmissionantennas is the equally spaced array arrangement of 12 elements. On theother hand, when no beam transmission antenna is used in the sameantenna arrangement as (a) in FIG. 20 and in a case (not illustrated)where the equally spaced arrangement is formed in the same manner as at(b) in FIG. 19, the virtual reception antenna arrangement is an equallyspaced array arrangement of eight elements since number Nt oftransmission antennas is 4 and number Na of reception antennas is 2.

As described above, it is possible to increase the number of beamtransmission antennas by increasing number N_(DM) of Dopplermultiplexing. It is thus possible to further increase the number ofvirtual reception antennas. Further, since encoder 107 uniformly setsthe numbers of coded Doppler multiplexing for the Doppler multiplexedsignals, all transmission antennas 109 are used for respective beamtransmission antennas in radar apparatus 10. Accordingly, the number ofbeam transmission antennas is easy to be increased in comparison withthe case where the numbers of coded Doppler multiplexing for the Dopplermultiplexed signal are set non-uniformly. In other words, the number oftransmission antennas 109 required to increase the number of beamtransmission antennas can be reduced.

For example, in order to set number N_(CM) of code multiplexing to 2 andnumber N_(BF) of beam transmission antennas to 1, number Nt oftransmission antennas>2 is requisite for the non-uniform setting of thenumbers of coded Doppler multiplexing for Doppler multiplexed signals.Meanwhile, number Nt of transmission antennas=2 is also applicable whenthe numbers of coded Doppler multiplexing for the Doppler multiplexedsignals are set uniformly.

An increase in the number of beam transmission antennas accompanyingsuch an increase in number N_(DM) of Doppler multiplexing allows forfurther enlargement of the aperture length of the virtual receptionantennas, and further improvement of the angular resolution in thevirtual reception antenna arrangement using the beam transmissionantenna. Further, by the virtual reception antennas densely arranged, itis possible to suppress the increase of the sidelobe, and to improve theangular resolution.

Note that, the example illustrated in FIG. 20 illustrates the case wherenumber N_(BF) of beam transmission antennas is 2, but number N_(BF) ofbeam transmission antennas is not limited thereto. For example, anincrease in the number of transmission antennas 109 allows for settingof a larger number of beam transmission antennas, thus improving theangular resolution of radar apparatus 10 or suppressing the sidelobelevel.

In addition, although FIG. 20 illustrates the case where a plurality oftransmission antennas 109 and reception antennas 202 are arrangedhorizontally, the arrangement of transmission antennas 109 and receptionantennas 202 is not limited thereto. For example, at least onetransmission antennas 109 or reception antennas 202 may be arrangedvertically, or may be arranged in a horizontal and vertical plane. Alsoin these cases, it is possible to achieve the same effect. Note that,the antennas illustrated in FIG. 20 may be a part of a plurality ofantennas that radar apparatus 10 includes.

Embodiment 2

The present embodiment will be described in relation to a method ofapplying a directivity weight to form the directivity of a beamtransmission antenna in a predetermined direction (a method ofcontrolling the directivity of the beam transmission antenna) inaddition to the operation of Embodiment 1.

[Configuration of Radar Apparatus]

In FIG. 21, components similar to those in Embodiment 1 (FIG. 1) areidentified with the same numerals, and a description thereof is omitted.

In radar transmitter 300 of radar apparatus 20, phase rotation amountsetter 105 may include directivity weight applier 301.

Directivity weight applier 301 outputs phase rotation amountsDW_(ndop_code(ndm_BF), ndm_BF)(M) to phase rotators 108. As given byfollowing Expression 69, each of phase rotation amountsDW_(ndop_code(ndm_BF), ndm_BF)(m) is obtained by further applying phaserotation DIR_(ndop_code(ndm_BF), ndm_BF)(θ) andTXCAL_(ndop_code(ndm_BF), ndm_BF) to ndm__(BF)-th coded Doppler phaserotation amount ψ_(ndop_code(ndm_BF), ndm_BF)(m) used for a beamtransmission antenna among coded Doppler phase rotation amountsψ_(ndop_code(ndm), ndm)(m) for m-th transmission period Tr that areinputted from encoder 107 and given by Expression 10.

$\begin{matrix}{\mspace{79mu}\lbrack 75\rbrack} & \; \\{{{DW}_{{{ndop\_ code}{({ndm\_ BF})}},{ndm\_ BF}}(m)} = {{\psi_{{{ndop\_ code}{({ndm\_ BF})}},{ndm\_ BF}}(m)} + {TxCAL}_{{{ndop\_ code}{({ndm\_ BF})}},{ndm\_ BF}} + {{DIR}_{{{ndop\_ code}{({ndm\_ BF})}},{ndm\_ BF}}(\theta)} - {{angle}\left\lbrack {{OC}_{{{ndop\_ code}{({ndm\_ BF})}},{ndm\_ BF}}\left( {noc}_{\_ BF} \right)} \right\rbrack}}} & \left( {{Expression}\mspace{14mu} 69} \right)\end{matrix}$

Here, the outputs of noc__(BF)-th Doppler analyzer 210 are inputted topeak extractor 213, with transmission period Tr in which OC_INDEX isnoc__(BF) being regarded as the transmission timing of the beamtransmission antenna. The character “noc_BF” represents an index of acode element corresponding to a timing (transmission period) at whichtransmission is performed by the beam transmission antenna. Here,noc__(BF) is any value of noc=1, . . . , Loc, which are indices ofN_(CM) (number of code multiplexing) code elements of orthogonal codesequences with code length Loc used in encoder 107.

In Expression 69, TxCAL_(ndop_code(ndm_BF), ndm_BF) is a phasecorrection factor for correcting a phase deviation (e.g., a phasedifference caused by a difference between the line lengths or pathlengths of antennas or power supply lines) betweenN_(DOP_CODE)(ndm__(BF)) transmission antennas of transmission antenna Tx#[1, ndm__(BF)], transmission antenna Tx #[2, ndm__(BF)], . . . , andtransmission antenna Tx #[N_(DOP_CODE)(ndm__(BF)), ndm__(BF)] forapplying ndm__(BF)-th Doppler shift DOP_(ndm_BF).

Further, in Expression 69, the term“angle[OC_(ndop_code(ndm_BF))(noc__(BF))]” is a phase correction factorfor forming a directivity weight by eliminating an influence of a phaserotation by code OC_(ndop_code(ndm_BF))(noc__(BF)) applied by encoder107 in transmission period Tr in which OC_INDEX is noc__(BF) (in otherwords, the phase rotation by code OC_(ndop_code(ndm_BF))(noc__(BF)) isused as the reference phase).

Further, in Expression 69, DIR_(ndop_code(ndm_BF), ndm_BF)(θ) is adirectivity weight factor for directing the directivity in apredetermined direction with respect to N_(DOP_CODE)(ndm__(BF))transmission antennas of transmission antenna Tx #[1, ndm__(BF)],transmission antenna Tx #[2, ndm__(BF)], . . . , and transmissionantenna Tx #[N_(DOP_CODE)(ndm__(BF)), ndm__(BF)] for applyingndm__(BF)-th Doppler shift DOP_(ndm_BF). In Expression 69, thedirectivity weight factor for directing the directivity in the azimuthdirection of azimuth θ is given as an example, but the presentdisclosure is not limited thereto. A directivity weight factor fordirecting the directivity direction in elevation angle direction φ orfor directing the directivity direction in the two dimensions includingazimuth θ and elevation angle direction φ may also be used.

Directivity weight factor DIR_(ndop_code(ndm_BF), ndm_BF)(θ) depends onthe arrangement of N_(DOP_CODE)(ndm__(BF)) transmission antennas oftransmission antenna Tx #[1, ndm__(BF)], transmission antenna Tx #[2,ndm__(BF)], . . . , transmission antenna Tx #[N_(DOP_CODE)(ndm__(BF)),ndm__(BF)] used as beam transmission antennas. For example, whenN_(DOP_CODE)(ndm__(BF)) transmission antennas 109 are linearly arrangedat element intervals d_(SA) and the transmission beam direction isdirected in the direction of θ_(TxBF), directivity weight applier 301generates directivity weight factorDIR_(ndop_code(ndm_BF), ndm_BF)(θ_(TxBF)) as given by followingExpression 70. Here, _(ndop_code(ndm_BF))=1, . . . ,N_(DOP_CODE)(ndm__(BF)).

$\begin{matrix}{\mspace{79mu}\lbrack 76\rbrack} & \; \\{{{{DIR}_{1,{ndm\_ BF}}\left( \theta_{TxBF} \right)} = 0},{{{DIR}_{2,{ndm\_ BF}}\left( \theta_{TxBF} \right)} = \frac{2\;\pi\; d_{SA}\sin\;\theta_{TxBF}}{\lambda}},{{{DIR}_{3,{ndm\_ BF}}\left( \theta_{TxBF} \right)} = \frac{4\;\pi\; d_{SA}\sin\;\theta_{TxBF}}{\lambda}},\ldots\mspace{14mu},{{{DIR}_{{N_{DOP\_ CODE}{({ndm\_ BF})}},{ndm\_ BF}}\left( \theta_{TxBF} \right)} = \frac{{2\;{\pi\left( {{N_{DOP\_ CODE}({ndm\_ BF})} - 1} \right)}d_{SA}\sin\;\theta_{TxBF}}\;}{\lambda}}} & \left( {{Expression}\mspace{14mu} 70} \right)\end{matrix}$

Here, λ denotes the wavelength of a radar transmission signal.

Note that, as given by following Expression 71, directivity weightapplier 301 may output, without any operation, ndm-th coded Dopplerphase rotation amount ψ_(ndop_code(ndm), ndm)(m) for m-th transmissionperiod Tr as given by Expression 10 that is inputted from encoder 107and that is not used for a beam transmission antenna, for example.

[77]

DW _(ndop_code(ndm),ndm)(m)=ψ_(ndop_code(ndm),ndm)(m)  (Expression 71)

In each transmission period Tr, each of Nt phase rotators 108 applies,to a chirp signal inputted from radar transmission signal generator 101,DW_(ndop_code(ndm), ndm)(m) inputted from directivity weight applier301. The Nt outputs (e.g., referred to as coded Doppler multiplexedsignals) of phase rotators 108 are radiated into space from Nttransmission antennas 109 (also referred to as “transmission arrayantenna section”) after amplified to a defined transmission power.

Radar receiver 200 of radar apparatus 20 illustrated in FIG. 21performs, for example, the same operation as that of Embodiment 1. Here,peak extractor 213 regards, as the transmission timing of the beamtransmission antenna, transmission period Tr in which OC_INDEX isnoc__(BF), and thus outputs, to direction estimator 214, the outputs ofnoc__(BF)-th Doppler analyzer 210 for distance index f_(b_cfar) andDoppler frequency index f_(s_comp_cfar) inputted from CFAR section 211.At this time, peak extractor 213 may use, for example, D_(rmin) that isa Doppler aliasing judgement result inputted from coded Dopplerdemultiplexer 212.

As described above, directivity weight applier 301 included in phaserotation amount setter 105 allows radar apparatus 20 to form thedirectivity of the beam transmission antenna in a predetermineddirection, so as to improve the directivity gain in the predetermineddirection. Thus, for example, when the viewing angle of radar apparatus20 is within about a 3-dB beam width of the beam transmission antenna,it is possible to extend the sensing range of radar apparatus 20.

Note that, when the viewing angle of radar apparatus 20 is within abouta 3-dB beam width of the beam transmission antenna, it is possible toextend the sensing range of radar apparatus 20. Thus, the output ofnoc__(BF) Doppler analyzer 210 may be used as the input to CFAR section211. For example, FIG. 22 illustrates an example in which, whennoc__(BF)=1 is used, the output of Doppler analyzer 210-1 is the inputto CFAR section 211. Note that noc__(BF) is not limited to 1.

In FIG. 22, CFAR section 211 calculates PowerFT(f_(b), f_(s)) obtainedby performing power addition of outputs VFT_(z) ^(noc_BF)(f_(b), f_(s))of noc__(BF)-th Doppler analyzer 210 as given by following Expression72. Subsequently, processing the same as that in the receptionprocessing of Embodiment 1 is performed.

[78]

PowerFT(f _(b) ,f _(s))=Σ_(z=1) ^(Na) |VFT _(z) ^(noc_BF)(f _(b) ,f_(s))|²  (Expression 72)

As illustrated in FIG. 22, when the viewing angle of radar apparatus 20is within about a 3-dB beam width of the beam transmission antenna, CFARsection 211 can perform CFAR processing using PowerFT(f_(b), f_(s))obtained by performing power addition of outputs VFT_(z)^(noc_BF)(f_(b), f_(s)) of noc__(BF) Doppler analyzer 210. It is thuspossible to improve the peak detection performance of CFAR section 211.For example, it is possible to improve the peak detection rate and toreduce the peak non-detection rate.

Further, for example, CFAR section 211 may perform power addition ofoutputs VFT_(z) ^(noc_BF)(f_(b), f_(s)) of noc__(BF)-th Doppler analyzer210, and it is thus possible to reduce the arithmetic processing amountof the power addition.

Note that, when a plurality of beam transmission antennas are present,the transmission timings (transmission periods) of a plurality of (e.g.,all) beam transmission antennas may be the same timing. For example, thetransmission timings of radar transmission signals may be the samebetween a plurality of beam transmission antennas (e.g., a first beamtransmission antenna and a second beam transmission antenna) eachconstituted by adjacent transmission antennas 109. It is thus possible,for example, to perform CFAR detection based on the outputs of oneDoppler analyzer 210 as illustrated in FIG. 22, to improve CFARprocessing performance and to reduce the arithmetic processing amount ofthe power addition in CFAR section 211.

For example, when there are a plurality of (e.g., two) beam transmissionantennas, directivity weight applier 301 further outputs phase rotationamounts DW_(ndop_code(ndm_BF1), ndm_BF1)(m) andDW_(ndop_code(ndm_BF2), ndm_BF2)(m) as given by following Expression 73to phase rotators 108 with respect to ndm__(BF1)-th and ndm__(BF2)-thcoded Doppler phase rotation amounts ψ_(ndop_code(ndm_BF1), ndm_BF1)(m)and ψ_(ndop_code(ndm_BF2), ndm_BF2)(m) among coded Doppler phaserotation amounts ψ_(ndop_code(ndm), ndm)(m) for m-th transmission periodTr inputted from encoder 107 and given by Expression 10. Here, a casewhere the outputs of noc__(BF)-th Doppler analyzer 210 are inputted topeak extractor 213, with transmission period Tr in which OC_INDEX isnoc__(BF) being regarded as the transmission timing of the two beamtransmission antennas is illustrated.

$\begin{matrix}{\mspace{79mu}\lbrack 79\rbrack} & \; \\{{{{DW}_{{{ndop\_ code}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}}(m)} = {{\psi_{{{ndop\_ code}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}}(m)} + {TxCAL}_{{{ndop\_ code}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}} + {{DIR}_{{{ndop\_ code}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}}(\theta)} - {{angle}\left\lbrack {{OC}_{{{ndop\_ code}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}}\left( {noc}_{BF} \right)} \right\rbrack}}}{{{DW}_{{{ndop\_ code}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}}(m)} = {{\psi_{{{ndop\_ code}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}}(m)} + {TxCAL}_{{{ndop\_ code}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}} + {{DIR}_{{{ndop\_ code}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}}(\theta)} - {{angle}\left\lbrack {{OC}_{{{ndop\_ code}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}}\left( {noc}_{BF} \right)} \right\rbrack}}}} & \left( {{Expression}\mspace{14mu} 73} \right)\end{matrix}$

Alternatively, when a plurality of beam transmission antennas arepresent, the transmission timings (transmission periods) of theplurality of (e.g., all) beam transmission antennas do not have to bethe same. For example, the transmission timings may be different betweena plurality of beam transmission antennas (e.g., a first beamtransmission antenna and a second beam transmission antenna) eachconstituted by adjacent transmission antennas 109. In this case, theCFAR detection based on the outputs of one Doppler analyzer 210 (e.g.,the configuration of FIG. 22) cannot be applied. However, since receivedpower variations according to transmission periods are leveled, thedynamic range of A/D can be narrowed. It is thus possible to obtain aneffect of reducibility of the number of AD quantization bits.

For example, when there are a plurality of (e.g., two) beam transmissionantennas, directivity weight applier 301 further outputs phase rotationamounts DW_(ndop_code(ndm_BF1), ndm_BF1)(m) andDW_(ndop_code(ndm_BF2), ndm_BF2)(m) as given by following Expression 74to phase rotators 108 with respect to ndm__(BF1)-th and ndm__(BF2)-thcoded Doppler phase rotation amounts ψ_(ndop_code(ndm_BF1), ndm_BF1)(m)and ψ_(ndop_code(ndm_BF2), ndm_BF2)(m) among coded Doppler phaserotation amounts ψ_(ndop_code(ndm), ndm)(m) for m-th transmission periodTr inputted from encoder 107 and given by Expression 10. Here, a case isillustrated in which the outputs of noc__(BF1)-th Doppler analyzer 210and noc__(BF2)-th Doppler analyzer 210 are inputted to peak extractor213, with transmission period Tr in which OC_INDEX is noc__(BF1) andtransmission period Tr in which OC_INDEX is noc__(BF2) being regarded asthe transmission timings of the two beam transmission antennas. Here,each of noc__(BF1) and noc__(BF2) is any value of noc=1, . . . , Loc,which are indices of N_(CM) (number of code multiplexing) code elementsof orthogonal code sequences with code length Loc used in encoder 107,and noc__(BF1)≠noc__(BF2) holds true.

$\begin{matrix}{\mspace{76mu}\lbrack 80\rbrack} & \; \\{{{D{W_{{{{ndop\_ cod}e}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}}(m)}} = {{\psi_{{{{ndop\_ cod}e}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}}(m)} + {TxCAL_{{{{ndop\_ cod}e}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}}} + {{DIR}_{{{{ndop\_ cod}e}{({{ndm\_ BF}\; 1})}},{{ndm\_ BF}\; 1}}(\theta)} - {{angle}\;\left\lbrack {{OC}_{ndop_{{cod{e{({ndm_{BF1}})}}},}ndm_{BF1}}\left( {noc}_{\_{BF}1} \right)} \right\rbrack}}}{{D{W_{{{{ndop\_ cod}e}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}}(m)}} = {{\psi_{{{{ndop\_ cod}e}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}}(m)} + {TxCAL_{{{{ndop\_ cod}e}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}}} + {{DIR}_{{{{ndop\_ cod}e}{({{ndm\_ BF}\; 2})}},{{ndm\_ BF}\; 2}}(\theta)} - {{angle}\;\left\lbrack {{OC}_{{{{ndop}\_{code}}{({{ndm}\_{BF}2})}},{{ndm}\_{BF}2}}\left( {noc}_{\_{BF}1} \right)} \right\rbrack}}}} & \left( {{Expression}\mspace{14mu} 74} \right)\end{matrix}$

In addition, when a plurality of beam transmission antennas are present,peak extractor 213 outputs, to direction estimator 214, at least one ofthe outputs of Doppler analyzers 210 for distance index f_(b_cfar) andDoppler frequency index f_(s_comp_cfar) inputted from CFAR section 211.At this time, peak extractor 213 may use D_(rmin) that is a Doppleraliasing judgement result inputted from coded Doppler demultiplexer 212.

Here, when a plurality of beam transmission antennas are present, andwhen the transmission periods serving as the plurality of (e.g., all)beam transmission antennas are to be the same, the outputs ofnoc__(BF)-th Doppler analyzer 210 are inputted to peak extractor 213,with transmission period Tr in which OC_INDEX is noc__(BF) beingregarded as the transmission timing of the beam transmission antennas,for example.

For example, when first (noc__(BF)=1) Doppler analyzer 210 is used, peakextractor 213 outputs

$\begin{matrix}\lbrack 81\rbrack & \; \\{{VF{T_{z}^{1}\left( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{ndm\_ BF}} \right)}}{N_{DM}}}} \right)}}.} & \;\end{matrix}$

Here, ndm__(BF) is any value of 1, . . . , N_(DM), and a plurality oftransmission antennas to which the ndm__(BF)-th Doppler multiplexedsignal is assigned satisfy the condition of the adjacent arrangement.

On the other hand, when a plurality of beam transmission antennas arepresent and the transmission periods serving as the beam transmissionantennas are not to be the same, the outputs from noc__(BF1)-th Doppleranalyzer 210 and noc__(BF2)-th Doppler analyzer 210 are inputted to peakextractor 213, for example, with transmission periods Tr in whichOC_INDEX is noc__(BF1) and in which OC_INDEX is noc__(BF2) beingregarded as the transmission timings of the beam transmission antennas.

Note that the operation performed when a plurality of beam transmissionantennas are present and the transmission timings of the beamtransmission antennas are to be the same or not to be the same is notlimited to the configuration according to the present embodiment (e.g.,the configuration for controlling the directivity of the beamtransmission antennas), but may also be applied, for example, to theconfiguration according to Embodiment 1 (e.g., the configuration inwhich the directivity control described above is not performed).

For example, FIG. 23 is a block diagram illustrating a configurationexample of radar apparatus 10 a in which an operation performed in thecase where the transmission periods serving as beam transmissionantennas are not to be the same when a plurality of beam transmissionantennas are present is adopted in Embodiment 1. In addition, FIG. 24 isa block diagram illustrating a configuration example of radar apparatus20 a in which an operation performed in the case where the transmissionperiods serving as beam transmission antennas are not to be the samewhen a plurality of beam transmission antennas are present is adopted inEmbodiment 2.

In FIGS. 23 and 24, for example, a case where ndm__(BF1)-th andndm__(BF2)-th coded Doppler multiplexed signals among coded Dopplermultiplexed signals are used as beam transmission antennas will bedescribed. For example, when noc__(BF1) corresponds to the transmissiontiming of ndm__(BF1)-th beam transmission antenna using first Doppleranalyzer 210 and noc__(BF2) corresponds to the transmission timing ofndm__(BF2)-th beam transmission antenna using second Doppler analyzer210, peak extractor 213 a outputs

$\begin{matrix}\lbrack 82\rbrack & \; \\{{{{VF}{T_{z}^{1}\left( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{ndm\_ BF1}} \right)}}{N_{DM}}}} \right)}},{and}}{{VF}{T_{z}^{2}\left( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{ndm\_ BF2}} \right)}}{N_{DM}}}} \right)}}} & \;\end{matrix}$

to direction estimator 214.

Here, ndm__(BF) is any value of 1, . . . , N_(DM), and a plurality oftransmission antennas to which the ndm__(BF)-th Doppler multiplexedsignal is assigned satisfy the condition of the adjacent arrangement.

In the present embodiment, directivity weight applier 301 may fix thedirectivity of a beam transmission antenna in a defined direction foreach of a plurality of measurements when forming the directivity in apredetermined direction, or may vary the directivity for eachmeasurement. When the directivity is variably set for each measurement,the directivity can be varied for each measurement by using directivityweight factor DIR_(ndop_code(ndm_BF), ndm_BF)(θ) in directivity weightapplier 301 that provides a θ direction different for each measurement.

[Antenna Arrangement Example 2-1]

Hereinafter, an example of antenna arrangement in the case ofnon-uniformly setting the numbers of coded Doppler multiplexing.Further, an example of antenna arrangement in the case where a pluralityof beam transmission antennas are used will be described.

For example, with reference to FIG. 25, a description will be given of acase where number Nt of transmission antennas used for multiplexingtransmission is 5, number N_(DM) of Doppler multiplexing is 3, N_(CM) is2, orthogonal code sequences Code₁ {1, 1} and Code₂ {1, −1} with codelength Loc=2 are set, and numbers N_(DOP_CODE)(1), N_(DOP_CODE)(2), andN_(DOP_CODE)(3) of coded Doppler multiplexing are 2, 2, and 1,respectively, in radar apparatus 20. Note that number N_(BF) of beamtransmission antennas is set to 2, and ndm__(BF1)=1 and ndm__(BF2)=2 areused as indices of Doppler multiplexed signals used for the beamtransmission antennas.

In FIG. 25, for example, horizontally arranged five transmissionantennas 109 (Tx #1 to Tx #5) are transmission antenna Tx #[1, 1],transmission antenna Tx #[2, 1], transmission antenna Tx #[1, 2],transmission antenna Tx #[2, 2], and transmission antenna Tx #[1, 3]from the left. In FIG. 25, two transmission antennas Tx #1 (Tx #[1, 1])and Tx #2 (Tx #[2, 1]) (first sub-array antenna) transmit radartransmission signals using the same Doppler multiplexing (Doppler shiftamount=DOP₁). Further, two adjacent transmission antennas Tx #3 (Tx #[1,2]) and Tx #4 (Tx #[2, 2]) (second sub-array antenna) transmit radartransmission signals using the same Doppler multiplexing (Doppler shiftamount=DOP₂). Thus, in FIG. 25, one beam transmission antenna is formedby Tx #1 and Tx #2, and one beam transmission antenna is formed by Tx #3and Tx #4. In FIG. 25, number N_(BF) of beam transmission antennas is 2.In the following, the beam transmission antenna based on Tx #1 and Tx #2may also be referred to as “Tx #6,” and the beam transmission antennabased on Tx #3 and Tx #4 may also be referred to as “Tx #7.”

Further, in FIG. 25, number Na of reception antennas is two (e.g., Rx #1and Rx #2). Note that, number Na of reception antennas is not limited totwo, and may be three or more, for example.

For example, when a radar transmission signal is transmitted fromadjacent Tx #1 (Tx #[1, 1]) and Tx #2 (Tx #[2, 1]), for example, at anequal power, the midpoint position between Tx #1 and Tx #2 serves as thephase center of beam transmission antenna Tx #6 (the cross markillustrated at (a) in FIG. 25). In addition, when a radar transmissionsignal is transmitted from adjacent Tx #3 (Tx #[1, 2]) and Tx #4 (Tx#[2, 2]), for example, at an equal power, the midpoint position betweenTx #3 and Tx #4 serves as the phase center of beam transmission antennaTx #7 (the cross mark illustrated at (a) in FIG. 25). Note that, whenthe radar transmission signal is not transmitted at an equal power fromtransmission antennas 109 constituting the beam transmission antennas,transmission at each position that is dependent on the ratio oftransmission powers of respective transmission antennas 109 constitutingthe beam transmission antenna (the position of the center of gravity ofthe transmission powers from the respective transmission antennas) andserves as the phase center of the sub-array can be treated astransmission by the beam transmission antenna.

Arrangement of VA #1 to VA #14 of virtual reception antennas (or MIMOvirtual antennas) as illustrated at (b) in FIG. 25 is constituted by thearrangement of transmission antennas Tx #1 to Tx #5, beam transmissionantennas Tx #6 and Tx #7, and reception antennas Rx #1 and Rx #2 asillustrated at (a) in FIG. 25. At (b) in FIG. 25, the virtual receptionantenna arrangement obtained based on beam transmission antenna Tx #6corresponds to VA #11 and VA #12, and the virtual reception antennaarrangement obtained based on beam transmission antenna Tx #7corresponds to VA #13 and VA #14.

Here, the arrangement of the virtual reception antennas (the virtualreception array) may be expressed by Expression 53, for example, basedon the positions of transmission antennas 109 constituting thetransmission array antenna (e.g., the positions of feeding points) andthe positions of reception antennas 202 constituting the reception arrayantenna (e.g., the positions of feeding points).

As illustrated at (b) in FIG. 25, since (Nt+N_(BF))=7 and Na=2, thevirtual reception antenna arrangement using the beam transmissionantennas is the equally spaced array arrangement of 14 elements. On theother hand, when no beam transmission antenna is used in the sameantenna arrangement as (a) in FIG. 25 and in a case (not illustrated)where the equally spaced arrangement is formed in the same manner as at(b) in FIG. 25, the virtual reception antenna arrangement is an equallyspaced array arrangement of 10 elements since number Nt of transmissionantennas is 5 and number Na of reception antennas is 2.

As is understood, an increase in number N_(DM) of Doppler multiplexingincreases the number of beam transmission antennas to allow for afurther increase in the number of virtual reception antennas. It is thuspossible for the virtual reception antenna arrangement using the beamtransmission antennas to further enlarge the aperture length and toimprove the angular resolution. Further, by the virtual receptionantennas densely arranged, it is possible to suppress the increase ofthe sidelobe, and to improve the angular resolution.

Note that, the example illustrated in FIG. 25 illustrates the case wherenumber N_(BF) of beam transmission antennas is two, but number N_(BF) ofbeam transmission antennas is not limited two. For example, an increasein the number of transmission antennas 109 allows for setting of alarger number of beam transmission antennas, thus improving the angularresolution of radar apparatus 10 or suppressing the sidelobe level.

In addition, although FIG. 25 illustrates the case where a plurality oftransmission antennas 109 and reception antennas 202 are arrangedhorizontally, the arrangement of transmission antennas 109 and receptionantennas 202 is not limited thereto. For example, at least onetransmission antennas 109 or reception antennas 202 may be arrangedvertically, or may be arranged in a horizontal and vertical plane. Alsoin these cases, it is possible to achieve the same effect. Note that,the antennas illustrated in FIG. 25 may be a part of a plurality ofantennas that radar apparatus 20 includes.

[Antenna Arrangement Example 2-2]

A two-dimensional antenna arrangement example using a sub-array will bedescribed. In addition, an example of antenna arrangement in the case ofuniformly setting the number of coded Doppler multiplexing will bedescribed.

For example, with reference to FIG. 26, a description will be given of acase where number Nt of transmission antennas used for multiplexingtransmission is 4, number N_(DM) of Doppler multiplexing is 2, N_(CM) is2, orthogonal code sequences Code₁ {1, 1} and Code₂ {1, −1} with codelength Loc=2 are set, and numbers N_(DOP_CODE)(1) and N_(DOP_CODE)(2) ofcoded Doppler multiplexing are 2 and 2 in radar apparatus 20. Note thatnumber N_(BF) of beam transmission antennas is set to 2, andndm__(BF1)=1 and ndm__(BF2)=2 are used as indices of Doppler multiplexedsignals used for the beam transmission antennas.

As illustrated in FIG. 26, a plurality of transmission antennas 109 andreception antennas 202 are arranged horizontally and vertically.

In FIG. 26, for example, transmission antennas 109 (Tx #1 and Tx #2)disposed in the vertically upper row are transmission antenna Tx #[1, 1]and transmission antenna Tx #[2, 1] from the left, and transmissionantennas 109 (Tx #3 and Tx #4) disposed in the vertically lower row aretransmission antenna Tx #[1, 2] and transmission antenna Tx #[2, 2] fromthe left.

In FIG. 26, two transmission antennas Tx #1 (Tx #[1, 1]) and Tx #2 (Tx#[2, 1]) (first sub-array antenna) transmit radar transmission signalsusing the same Doppler multiplexing (Doppler shift amount=DOP₁).Further, in FIG. 26, two adjacent transmission antennas Tx #3 (Tx #[1,2]) and Tx #4 (Tx #[2, 2]) (second sub-array antenna) transmit radartransmission signals using the same Doppler multiplexing (Doppler shiftamount=DOP₂). Thus, in FIG. 26, one beam transmission antenna is formedby Tx #1 and Tx #2, and one beam transmission antenna is formed by Tx #3and Tx #4. In FIG. 26, number N_(BF) of beam transmission antennas is 2.In the following, the beam transmission antenna based on Tx #1 and Tx #2may also be referred to as “Tx #5,” and the beam transmission antennabased on Tx #3 and Tx #4 may also be referred to as “Tx #6.”

Further, in FIG. 26, number Na of reception antennas is eight (e.g., Rx#1 to #8). Note that, number Na of reception antennas is not limited toeight, and any number of reception antennas may be present.

For example, when a radar transmission signal is transmitted fromhorizontally adjacent Tx #1 (Tx #[1, 1]) and Tx #2 (Tx #[2, 1]), forexample, at an equal power, the midpoint position between Tx #1 and Tx#2 serves as the phase center of beam transmission antenna Tx #5 (thecross mark illustrated at (a) in FIG. 26). In addition, when a radartransmission signal is transmitted from horizontally adjacent Tx #3 (Tx#[1, 2]) and Tx #4 (Tx #[2, 2]), for example, at an equal power, themidpoint position between Tx #3 and Tx #4 serves as the phase center ofbeam transmission antenna Tx #6 (the cross mark illustrated at (a) inFIG. 26). Note that, when the radar transmission signal is nottransmitted at an equal power from transmission antennas 109constituting the beam transmission antennas, transmission at a positiondependent on the ratio of transmission powers of respective transmissionantennas 109 constituting the beam transmission antenna (the position ofthe center of gravity of the transmission powers from the respectivetransmission antennas) that serves as the phase center of the sub-arraycan be treated as transmission by the beam transmission antenna.

Further, an antenna having a sub-array configuration as illustrated, forexample, in FIG. 27 may be used for transmission antenna 109. By usingthe antenna having the sub-array configuration, it is possible toimprove the directivity gain of the antenna to improve the sensingperformance (e.g., sensing range) of radar apparatus 20. For example, inthe example illustrated in FIG. 27, each of four transmission antennas109 (Tx #1 to Tx #4) has the sub-array configuration with six elementsof three vertically-arranged planar patch antennas and twohorizontally-arranged planar patch antennas. Note that the sub-arrayconfiguration is not limited to the configuration illustrated in FIG.27.

Arrangement of VA #1 to VA #48 of virtual reception antennas (or MIMOvirtual antennas) as illustrated in FIG. 28 is constituted by thearrangement of transmission antennas Tx #1 to Tx #4, beam transmissionantennas Tx #5 and Tx #6, and reception antennas Rx #1 to Rx #8 asillustrated in FIG. 26. In FIG. 28, the virtual reception antennaarrangement obtained based on beam transmission antenna Tx #5corresponds to VA #33 to VA #40, and the virtual reception antennaarrangement obtained based on beam transmission antenna Tx #6corresponds to VA #41 to VA #48.

Here, the arrangement of the virtual reception antennas (the virtualreception array) may be expressed by Expression 53, for example, basedon the positions of transmission antennas 109 constituting thetransmission array antenna (e.g., the positions of feeding points) andthe positions of reception antennas 202 constituting the reception arrayantenna (e.g., the positions of feeding points).

As illustrated FIG. 28, since (Nt+N_(BF))=6 and Na=8, the virtualreception antenna arrangement using the beam transmission antennas isthe equally spaced array arrangement of 48 elements. On the other hand,the virtual reception antenna arrangement in a case where no beamtransmission antenna is used in the same antenna arrangement as FIG. 26is the array arrangement of 32 elements since number Nt of transmissionantennas is 4 and number Na of reception antennas is 8, for example, asillustrated in FIG. 29.

As is understood, an increase in number N_(DM) of Doppler multiplexingincreases the number of beam transmission antennas to allow for afurther increase in the number of virtual reception antennas. It is thuspossible for the virtual reception antenna arrangement using the beamtransmission antennas to further enlarge the aperture length and toimprove the angular resolution. Further, by the virtual receptionantennas densely arranged, it is possible to suppress the increase ofthe sidelobe, and to improve the angular resolution.

Note that, the example illustrated in FIG. 26 illustrates the case wherenumber N_(BF) of beam transmission antennas is two, but number N_(BF) ofbeam transmission antennas is not limited two. For example, an increasein the number of transmission antennas 109 allows for setting of alarger number of beam transmission antennas, thus improving the angularresolution of radar apparatus 10 or suppressing the sidelobe level.

The antenna arrangement example has been described above.

Parts (a) and (b) in FIG. 30 illustrate examples of direction estimationresults (computer simulation results) obtained using the beamformermethod for the direction-of-arrival estimation algorithm of directionestimator 214.

Parts (a) and (b) in FIG. 30 illustrate plotted outputs of adirection-of-arrival estimation evaluation function value. The outputswere obtained within the range of ±90 degrees in the horizontaldirection and within the range of ±90 degrees in the vertical directionon the assumption that the target true value is horizontal 0 degrees andvertical 0 degrees. Note that, calculation was performed assuming thethe directivity of each antenna is omni-directivity.

For example, part (a) in FIG. 30 illustrates an example of the directionestimation result obtained by using the 48-element virtual receptionantenna arrangement (where D_(H)=0.5λ and D_(V)=0.5λ) using the beamtransmission antennas illustrated in FIG. 28. In addition, forcomparison with (a) in FIG. 30, (b) in FIG. 30 illustrates an example ofthe direction estimation result obtained by using the 32-element virtualreception antenna arrangement (where D_(H)=0.5λ and D_(V) is 0.5λ)constituted by number Nt=4 of transmission antennas and number Na=8 ofreception antennas illustrated in FIG. 29.

Part (b) in FIG. 30 illustrates that sidelobes were generatedhorizontally and vertically in other directions than the direction ofthe horizontal 0 degrees and vertical 0 degrees of the target truevalue. For example, (b) in FIG. 30 illustrates that more sidelobes weregenerated horizontally. In contrast, with reference to (a) in FIG. 30 ascompared with (b) in FIG. 30, it can be confirmed that the peak levelsof the sidelobes (e.g., horizontal sidelobes) in the other directionsthan the direction of the horizontal 0 degrees and vertical 0 degrees ofthe target true value were reduced. For example, the ratio (peak tosidelobe ratio (PSLR)) of the peak power value of the highest horizontalsidelobe to the peak power value of the main lobe in the direction ofthe horizontal 0 degrees and vertical 0 degrees was about 13 dB in thecase of (a) in FIG. 30, whereas the PSLR of the horizontal sidelobe wasabout 5 dB in the case of (b) in FIG. 30. The highest horizontalsidelobe is the highest except for the main lobe among the sidelobes inthe other directions than the direction of the horizontal 0 degrees andvertical 0 degrees. Therefore, it can be confirmed that a PSLR reductioneffect is higher in the case of (a) in FIG. 30 than in the case of (b)in FIG. 30.

As is understood, even when an element size in a MIMO array arrangementillustrated in FIG. 26 (or FIG. 27) is about 1k, the use of the beamtransmission antenna makes it possible to obtain a reduction effect ofreducing a horizontal grating lobe or sidelobe in the virtual receptionantenna. Note that, the antennas illustrated in FIGS. 26 and 27 may bepart of a plurality of antennas that radar apparatus 20 includes.

The embodiments of the present disclosure have been described above.

Other Embodiments

(1) The above-described embodiments have been described in relation tothe operation in which Doppler shift setter 106 sets phase rotationamount φ_(ndm) for applying Doppler shift amount DOP_(ndm) for each oftransmission periods (e.g., Loc×Tr transmission periods) correspondingto code length Loc of orthogonal code sequences, for example, andoutputs phase rotation amount φ_(ndm) to encoder 107. For example,Doppler shift setter 106 may set Doppler shift amounts DOP₁, DOP₂, . . ., and DOP_(N_DM) variably for respective code elements of the orthogonalcode sequences with code length Loc used for code multiplexing. In otherwords, the phase rotation amounts for applying Doppler shift amountsDOP₁, DOP₂, . . . , and DOP_(N_DM) may be set variably for respectivecode elements of the orthogonal code sequences with code length Loc usedfor code multiplexing.

For example, Doppler shift setter 106 may assign different phaserotation amount φ_(ndm)(noc) for each of transmission periods (Loc×Trtransmission periods) for the code element. Here, noc denotes the indexof a code element, and noc=1, . . . , Loc. In other words, in the m-thtransmission period, phase rotation amount φ_(ndm)(OC_INDEX) may beapplied variably depending on OC_INDEX=mod(m−1, Loc)+1. Here OC_INDEX=1,. . . , Loc. For example, phase rotation amount φ_(ndm)(noc)corresponding to the same Doppler shift amount DOP_(ndm) differs foreach transmission period corresponding to the code element of theorthogonal code sequences. Here, ndm=1, . . . , N_(DM). In other words,for example, Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM)differ between Tx #[1, 1] and Tx #[2, 1] illustrated in FIG. 12 forrespective code elements of the orthogonal code sequences.

For example, Doppler shift setter 106 may set Doppler shift amountsDOP₁, DOP₂, and DOP_(N_DM) differently for respective code elements ofthe orthogonal code sequences even when the pair of adjacenttransmission antennas as illustrated in FIG. 12 is not associated withcombinations of the same Doppler shift amount.

When phase rotators 108 set, as the same phase rotation amount (the sameDoppler shift amount) for a pair of adjacent transmission antennas 109,the Doppler shift amount set by Doppler shift setter 106, encoder 107may set the same number of coded Doppler multiplexing or may setdifferent numbers of coded Doppler multiplexing.

Examples of a method of variably setting the phase rotation amounts forapplying Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM) forrespective code elements of the orthogonal codes with code length Locused for code multiplexing include the following three methods.

<Phase Rotation Amount Varying Method 1>

Doppler shift setter 106 may set phase rotation amounts variably for thephase rotation amounts for applying Doppler shift amounts DOP₁, DOP₂, .. . , and DOP_(N_DM), for example, based on the equal-interval Dopplershift amount setting of intervals narrower than the intervals of themaximum equal-interval Doppler shift amount setting and based on themaximum equal-interval Doppler shift amount setting. In other words,Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM) may be variablyset for respective code elements of the orthogonal code sequences.

For example, when number N_(CM)=2 of code multiplexing is used (in thecase of code length Loc=2), Doppler shift setter 106 may variably setthe phase rotation amounts for applying Doppler shift amounts DOP₁,DOP₂, . . . , and DOP_(N_DM) based on the equal-interval Doppler shiftamount setting (e.g., Expression 6) of intervals narrower than theintervals of the maximum equal-interval Doppler shift amount setting inthe case of noc=1 (or in a transmission period in which OC_INDEX=1), orbased on the maximum equal-interval Doppler shift amount setting (e.g.,Expression 5) in the case of noc=2 (or in a transmission period in whichOC_INDEX=2). In other words, Doppler shift amounts DOP₁, DOP₂, . . . ,and DOP_(N_DM) may be set variably for respective code elements of theorthogonal code sequences. In this instance, phase rotation amountφ_(ndm)(noc) of the code element for each of the transmission periods(Loc×Tr transmission periods) is expressed by, for example, followingExpressions 75 and 76:

$\begin{matrix}\lbrack 83\rbrack & \; \\{{{\phi_{ndm}(1)} = \frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM} + N_{int}}};} & \left( {{Expression}\mspace{14mu} 75} \right) \\\lbrack 84\rbrack & \; \\{{\phi_{ndm}(2)} = {\frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM}}.}} & \left( {{Expression}\mspace{14mu} 76} \right)\end{matrix}$

<Phase Rotation Amount Varying Method 2>

Doppler shift setter 106 may use phase rotation amounts with variablyset N_(int) for the phase rotation amounts for applying Doppler shiftamounts DOP₁, DOP₂, . . . , and DOP_(N_DM), for example, when beingbased on the equal-interval Doppler shift amount setting of intervalsnarrower than the intervals of the maximum equal-interval Doppler shiftamount setting. In other words, Doppler shift amounts DOP₁, DOP₂, . . ., and DOP_(N_DM) may be variably set for respective code elements of theorthogonal code sequences.

For example, when number N_(CM)=2 of code multiplexing is used (in thecase of code length Loc=2), Doppler shift setter 106 may variably setthe phase rotation amounts for applying Doppler shift amounts DOP₁,DOP₂, . . . , and DOP_(N_DM) based on N_(int)=1 in the equal-intervalDoppler shift amount setting (e.g., Expression 6) in the case of noc=1(or in a transmission period in which OC_INDEX=1), or based on N_(int)=2in the equal-interval Doppler shift amount setting (e.g., Expression 6)in the case of noc=2 (or in a transmission period in which OC_INDEX=2).In other words, Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM)may be variably set for respective code elements of the orthogonal codesequences. In this instance, phase rotation amount φ_(ndm)(noc) of thecode element for each of the transmission periods (Loc×Tr transmissionperiods) is expressed by, for example, following Expressions 77 and 78:

$\begin{matrix}\lbrack 85\rbrack & \; \\{{{\phi_{ndm}(1)} = \frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM} + 1}};} & \left( {{Expression}\mspace{14mu} 77} \right) \\\lbrack 86\rbrack & \; \\{{\phi_{ndm}(2)} = {\frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM} + 2}.}} & \left( {{Expression}\mspace{14mu} 78} \right)\end{matrix}$

<Phase Rotation Amount Varying Method 3>

Doppler shift setter 106 may variably set indices of the phase rotationamounts for applying Doppler shift amounts DOP₁, DOP₂, . . . , andDOP_(N_DM), for example, when being based on the equal-interval Dopplershift amount setting of intervals narrower than the intervals of themaximum equal-interval Doppler shift amount setting. In other words,Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM) may be variablyset for respective code elements of the orthogonal code sequences.

For example, when number NCM=2 of code multiplexing is used (in the caseof code length Loc=2), phase rotation amount φ_(ndm)(noc) for the codeelement for each of the transmission periods (Loc×Tr transmissionperiods) is expressed by following Expressions 79 and 80. Here, noc=1, .. . , Loc.

$\begin{matrix}\lbrack 87\rbrack & \; \\{{\phi_{ndm}(1)} = \frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM} + N_{int}}} & \left( {{Expression}\mspace{14mu} 79} \right) \\\lbrack 88\rbrack & \; \\{{\phi_{ndm}(2)} = \frac{2{\pi({ndm})}}{N_{DM} + N_{int}}} & \left( {{Expression}\mspace{14mu} 80} \right)\end{matrix}$

In the case of using Expressions 79 and 80, an index setting for theDoppler shift amount for the transmission period in which a first codeelement is transmitted is an index setting with an index (ndm) shiftedby one index as compared with an index setting for the Doppler shiftamount for the transmission period in which a second code element istransmitted.

Note that, a method of index shifting is not limited to the above, andother methods for shifting may also be used. For example, when themethod of index shifting is known in advance, separation processing canbe performed by coded Doppler demultiplexer 212 of radar receiver 200.

The example of the method of variably setting the phase rotation amounts(in other words, the method of variably setting the Doppler shiftamounts) has been described above.

Hereinbelow, by way of example, a description will be given of adifferent part of the operation different between the present embodimentand Embodiment 1 regarding the operation performed in the case where thephase rotation amounts for applying Doppler shift amounts DOP₁, DOP₂, .. . , and DOP_(N_DM) are set variably for respective code elements ofthe orthogonal code sequences with code length Loc used for codemultiplexing (in other words, in the case where Doppler shift amountsDOP₁, DOP₂, . . . , and DOP_(N_DM) are set variably for respective codeelements of the orthogonal code sequences), e.g., in the case wherephase rotation amounts φ_(ndm)(noc) corresponding respectively to thecode elements and being different from one another for each of thetransmission periods (Loc×Tr transmission periods) are applied.

Note that, in the following, application of phase rotation amountsφ_(ndm)(noc) is denoted by N_(int)(noc). For example, Expressions 75 and76 in phase rotation amount varying method 1 are expressed as followingExpressions 81 and 82:

$\begin{matrix}\lbrack 89\rbrack & \; \\{{{\phi_{ndm}(1)} = \frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM} + {N_{int}(1)}}},{{{N_{int}(1)} = 1};}} & \left( {{Expression}\mspace{14mu} 81} \right) \\\lbrack 90\rbrack & \; \\{{{\phi_{ndm}(2)} = \frac{2{\pi\left( {{ndm} - 1} \right)}}{N_{DM} + {N_{int}(2)}}},{{N_{int}(2)} = 0.}} & \left( {{Expression}\mspace{14mu} 82} \right)\end{matrix}$

For example, with respect to the phase rotation amounts for applyingN_(DM) Doppler shift amounts inputted from Doppler shift setter 106,encoder 107 sets the phase rotation amounts based on the orthogonal codesequences. For example, with respect to phase rotation amountφ_(ndm)(OC_INDEX) for applying ndm-th Doppler shift amount DOP_(ndm),encoder 107 may set coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm)(m) for m-th transmission period Tr that is givennot by Expression 10 but by following Expression 83, and may outputcoded Doppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m) to phaserotator 108. Here, ndop_code(ndm)=1, . . . , N_(DOP_CODE)(ndm) andndm=1, . . . , N_(DM).

$\begin{matrix}{\mspace{79mu}\lbrack 91\rbrack} & \; \\{{\Psi_{{ndop_{-}cod{e{({ndm})}}},{ndm}}(m)} = {{{{floor}\mspace{11mu}\left\lbrack \frac{m - 1}{Loc} \right\rbrack} \times {\phi_{ndm}({OC\_ INDEX})}} + {{angle}\;\left\lbrack {{OC}_{{ndop\_ code}{({ndm})}}({OC\_ INDEX})} \right\rbrack}}} & \left( {{Expression}\mspace{14mu} 83} \right)\end{matrix}$

Subsequent operations of radar transmitter 100 are the same as those ofEmbodiment 1.

Next, a difference in the operation of radar receiver 200 between thepresent embodiment and Embodiment 1 will be described.

Since the equal-interval Doppler shift amount setting of intervalsnarrower than the intervals of the maximum equal-interval Doppler shiftamount setting is applied in all of phase rotation amount varyingmethods 1 to 3, CFAR section 211 may perform the following processing.

For example, when phase rotation amount varying method 1 is used, CFARsection 211 performs power addition of some of outputs VFT_(z)^(noc)(f_(b), f_(s)) of Doppler analyzers 210 of first to Na-th signalprocessors 206 based on noc (or transmission period in whichOC_INDEX=noc) corresponding to the setting of the phase rotation amountbased on the equal-interval Doppler shift amount setting of intervalsnarrower than the intervals of the maximum equal-interval Doppler shiftamount setting. For example, CFAR section 211 may perform power additionusing outputs VFT_(z) ^(noc)(f_(b), f_(s)) of Doppler analyzer 210 equalto noc (or the transmission period in which OC_INDEX=noc) correspondingto the setting of the phase rotation amount based on the equal-intervalDoppler shift amount setting of intervals narrower than the intervals ofthe maximum equal-interval Doppler shift amount setting.

For example, in the case of number NCM of code multiplexing is 2, CFARsection 211 may perform power addition of outputs VFT_(z) ¹(f_(b),f_(s)) of Doppler analyzers 210 of first to Na-th signal processors 206as given by following Expression 84 instead of by Expression 38 whenExpression 6 is used as the equal-interval Doppler shift amount settingof intervals narrower than the intervals of the maximum equal-intervalDoppler shift amount setting in the case of noc=1 (or the transmissionperiod in which OC_INDEX=1), or when Expression 5 is used as the maximumequal-interval Doppler shift amount setting in the case of noc=2 (or thetransmission period in which OC_INDEX=2):

[92]

PowerFT(f _(b) ,f _(s))=Σ_(z=1) ^(Na) |VFT _(z) ¹(f _(b) ,f_(s))|²  (Expression 84).

In addition, CFAR section 211 may perform two-dimensional CFARprocessing in two dimensions formed by a distance axis and a Dopplerfrequency axis (corresponding to relative velocity), or CFAR processingusing one-dimensional CFAR processing in combination, for example, basedon a power addition value.

Further, for example, when phase rotation amount varying method 2 isused, CFAR section 211 performs the power addition of some of outputsVFT_(z) ^(noc)(f_(b), f_(s)) of Doppler analyzers 210 of first to Na-thsignal processors 206 using one noc (or the transmission period in whichOC_INDEX=noc). For example, when Expression 6 is used, CFAR section 211may perform the power addition using noc (or the transmission period inwhich OC_INDEX=noc) corresponding to the phase rotation amount settingwith minimum N_(int). For example, when Expression 6 is used, CFARsection 211 may perform the power addition using outputs VFT_(z)^(noc)(f_(b), f_(s)) of Doppler analyzer 210 with noc that is equal tonoc corresponding to the phase rotation amount setting with minimumN_(int) (or the transmission period in which OC_INDEX=noc).

For example, when noc=1 (or the transmission period in which OC_INDEX=1)corresponds to the phase rotation amount setting with minimum N_(int) inExpression 6 (hereinafter, N_(int) in this case is referred to asN_(intMIN)), CFAR section 211 may perform the power addition of outputsVFT_(z) ¹(f_(b), f_(s)) of Doppler analyzers 210 of first to Na-thsignal processors 206 as given by following Expression 85 instead ofExpression 38:

[93]

PowerFT(f _(b) ,f _(s))=Σ_(z=1) ^(Na) |VFT _(z) ¹(f _(b) ,f_(s))|²  (Expression 85).

In addition, CFAR section 211 may perform two-dimensional CFARprocessing in two dimensions formed by a distance axis and a Dopplerfrequency axis (corresponding to relative velocity), or CFAR processingusing one-dimensional CFAR processing in combination, for example, basedon a power addition value.

For power addition value PowerFT(f_(b), f_(s)) used in the CFARprocessing in any of above-mentioned phase rotation amount varyingmethods 1 to 3, the equal-interval Doppler shift amount setting usingphase rotation amount φ_(ndm) given by Expression 6 is used as theequal-interval Doppler shift amount setting of intervals narrower thanthe intervals of the maximum equal-interval Doppler shift amountsetting. Accordingly, N_(DM) peaks are detected at intervals ofΔFD=Ncode/(N_(DM)+N_(int)), and it is thus possible for CFAR section 211to apply the Doppler domain compression CFAR processing.

For example, as given by following Expression 86, CFAR section 211 mayperform the power addition while adjusting the peak positions of Dopplershift multiplexed signals to perform the Doppler domain compression CFARprocessing. Here, f_(s_comp)=−ΔFD/2, . . . ,ΔFD/2−1=Ncode/{2(N_(DM)+N_(int))}, . . . , Ncode/{2(N_(DM)+N_(int))}−1.

[94]

Phase rotation amount varying method 1 or 3:

$\begin{matrix}{{PowerFT_{com{p{({f_{b},f_{s\_ comp}})}}}} = {\sum\limits_{{nfd} = 1}^{N_{DM} + N_{int}}{{PowerFT}\left( {f_{b},{f_{s\_ comp}\  + {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right) \times \Delta FD}}} \right)}}} & \left( {{Expression}\mspace{14mu} 86} \right)\end{matrix}$

Note that, in the following, “N_(intMIN)” is used instead of N_(int) inthe expressions and descriptions when phase rotation amount varyingmethod 2 is used.

However, in Expression 86, in the case of

$\begin{matrix}\lbrack 95\rbrack & \; \\{{{f_{s\_ comp}\  + {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right) \times \Delta\;{FD}}} < {{- {Ncode}}\text{/}2}},} & \;\end{matrix}$

the Doppler frequency index to which Ncode is added is used.

Likewise, in Expression 86, in the case of

$\begin{matrix}\lbrack 96\rbrack & \; \\{{{f_{s\_ comp}\  + {\left( {{nfd} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right) \times \Delta\;{FD}}} > {\frac{N_{code}}{2} - 1}},} & \;\end{matrix}$

the Doppler frequency index from which Ncode is further subtracted isused.

It is thus possible to compress the Doppler frequency range for the CFARprocessing to 1/(N_(DM)+N_(int)) to reduce the amount of CFAR processingand to simplify the circuit configuration. In addition, CFAR section 211is capable of power addition for N_(DM) Doppler-shift multiplexedsignals, to improve SNR by about (N_(DM))^(1/2). As a result, the radarsensing performance of radar apparatus 10 can be improved.

CFAR section 211 using the Doppler domain compression CFAR processingadaptively sets the threshold, for example, and outputs, to codedDoppler demultiplexer 212, distance index f_(b_cfar) and Dopplerfrequency index f_(s_comp_cfar) that provide a received power greaterthan the threshold, and received power information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD (where nfd=1, . . ., N_(DM)+N_(int))) for the Doppler frequency indices(f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD of N_(DM) Dopplermultiplexed signals.

The exemplary operation of CFAR section 211 has been described above.

Next, an example of the operation of coded Doppler demultiplexer 212illustrated in FIG. 1 will be described. The following describes anexample of processing performed by coded Doppler demultiplexer 212 whenCFAR section 211 uses the Doppler domain compression CFAR processing.

Based on the outputs of CFAR section 211 (e.g., distance indicesf_(b_cfar), Doppler frequency indices f_(s_comp_cfar), and receivedpower information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD (where nfd=1, . . ., N_(DM)+N_(int))) for the Doppler frequency indices(f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) of(N_(DM)+N_(int)) Doppler multiplexed signals), coded Dopplerdemultiplexer 212 separates the coded Doppler multiplexed transmissionsignals using the outputs of Doppler analyzers 210, and distinguishes(in other words, judges or identifies) transmission antennas 109 and theDoppler frequencies (in other words, the Doppler velocities or relativevelocities).

As described above, in the case where the numbers of coded Dopplermultiplexing for the Doppler multiplexed signals are set by using theequal-interval Doppler shift amount setting of intervals narrower thanthe intervals of the maximum equal-interval Doppler shift amount setting(e.g., Expression 6), coded Doppler demultiplexer 212 performs, forexample, (1) the aliasing judgement, and (2) the Doppler code separationprocessing on the coded Doppler multiplexed signals used formultiplexing transmission based on the result of the aliasing judgement.

Processing (1) and processing (2) by coded Doppler demultiplexer 212described above will be described below.

<(1) Aliasing Judgement Processing (Detection Processing of DetectingUnused Coded Doppler Multiplexed Signal)>

For example, in the aliasing judgement, coded Doppler demultiplexer 212detects N_(DM) peaks at intervals of ΔFD=Ncode/(N_(DM)+N_(int)). Forexample, coded Doppler demultiplexer 212 detects N_(DM) peaks atintervals of ΔFD using the outputs of CFAR section 211 (e.g., distanceindices f_(b_cfar), Doppler frequency indices f_(s_comp_cfar), andreceived power information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD (where nfd=1, . . ., N_(DM)+N_(int))) for the Doppler frequency indices(f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) of(N_(DM)+N_(int)) Doppler multiplexed signals). For example, usingreceived power information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) for Dopplerfrequency indices (f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD)of the (N_(DM)+N_(int)) Doppler multiplexed signals, coded Dopplerdemultiplexer 212 detects the Doppler frequency indices of N_(int) codedDoppler multiplexed signals not used for multiplexing transmission.Through this processing, coded Doppler demultiplexer 212 performs thealiasing judgement in the Doppler range of ±1/(2Loc×Tr).

Here, the detection of the Doppler frequency indices of N_(int) codedDoppler multiplexed signals not used for Doppler multiplexingtransmission may be performed using received power informationPowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) as follows.

For example, when N_(int)=1, coded Doppler demultiplexer 212 detects aD_(r) in which received power PowerFT(f_(b_cfar),f_(s_comp_cfar)+(D_(r)−ceil((N_(DM)+N_(int))/2)−1)×ΔFD) is minimizedamong the D_(r) ranges, as given by following Expression 87. Such aD_(r) is expressed as “D_(r min).” Here, D_(r) is an integer value in arange of D_(r)=−ceil((N_(DM)+N_(int))/2), . . . ,ceil((N_(DM)+N_(int))/2)−1.

$\begin{matrix}{\mspace{79mu}\lbrack 97\rbrack} & \; \\{D_{rmin} = \left\{ {{\arg\; D_{r}}❘{\min\limits_{D_{r}}\;{{PowerFT}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \ {\left( {D_{r} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right)\Delta\;{FD}}}} \right)}}} \right\}} & \left( {{Expression}\mspace{14mu} 87} \right)\end{matrix}$

For example, when N_(int)>2, coded Doppler demultiplexer 212 detects theD_(r) of minimized power by utilizing the beforehand knowledge ofrelative positional relationship between the Doppler frequency indicesof N_(int) coded Doppler multiplexed signals not used for Dopplermultiplexing transmission in respective D_(r). For example, whenN_(int)>2, coded Doppler demultiplexer 212 detects a D_(r) in which thereceived power is minimized among the D_(r) using following Expression88. Such a D_(r) is expressed as “D_(r min).” Here, D_(r) is an integervalue in a range of D_(r)=−ceil((N_(DM)+N_(int))/2), . . . ,ceil((N_(DM)+N_(int))/2)−1. Here, F_(nint)(D_(r)) is an indexrepresenting the relative positional relation of the Doppler frequencyindex of the nint-th coded Doppler multiplexed signal not used forDoppler multiplexing transmission in D_(r). Note that the index intervalbetween that indices represented by F_(nint)(D_(r)) is ΔFD. Here,nint=1, . . . , N_(int).

$\begin{matrix}{\mspace{79mu}\lbrack 98\rbrack} & \; \\{D_{rmin} = \left\{ {{\arg\; D_{r}}❘{\min\limits_{D_{r}}{\sum\limits_{{nint} = 1}^{N_{int}}{{PowerFT}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \ \left( {{F_{nint}\left( D_{r} \right)} - {{ceil}\ \left( \frac{N_{DM} + N_{int}}{2} \right)} - 1} \right) + {\Delta\;{FD}}}} \right)}}}} \right\}} & \left( {{Expression}\mspace{14mu} 88} \right)\end{matrix}$

Coded Doppler demultiplexer 212 outputs, to peak extractor 213, analiasing judgement result (e.g., f_(b_cfar), f_(s_comp_cfar), D_(rmin)),for example, with respect to a reception signal for f_(b_cfar) andf_(s_comp_cfar).

The operation example of the aliasing processing has been describedabove.

<(2) Doppler Code Separation Processing on Coded Doppler MultiplexedSignal Used for Multiplexing Transmission>

Coded Doppler demultiplexer 212 performs coded Doppler demultiplexingprocessing on a coded Doppler multiplexed signal used for multiplexingtransmission based on an aliasing judgement result.

For example, coded Doppler demultiplexer 212 applies Expression 51 basedon D_(rmin) which is a result of aliasing judgement in aliasingjudgement processing, so as to separate and receive the coded Dopplermultiplexed signal to which DCI (ncm, ndm) used for multiplexingtransmission is assigned. For example, coded Doppler demultiplexer 212can perform the separation processing using Expression 51 to separateand receive the coded Doppler multiplexed signal to which DCI (ncm, ndm)used for the multiplexing transmission is assigned.

Note that, following Expression 89 may also be used forVFTALL_(z)(f_(b_cfar), f_(s_comp_cfar), D_(r), ndm) in Expression 51.

$\begin{matrix}{\mspace{79mu}\lbrack 99\rbrack} & \; \\{{{VFTALL}_{z}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r},{ndm}} \right)} = \left\lbrack {{{VFT}_{z}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r},{ndm},1} \right)}}{N_{DM} + {N_{int}(1)}}}} \right)}\mspace{14mu}\ldots\mspace{14mu}{{VFT}_{z}^{L_{oc}}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r},{ndm},L_{oc}} \right)}}{N_{DM} + {N_{int}\left( L_{oc} \right)}}}} \right\rbrack}^{T}} \right.} & \left( {{Expression}\mspace{14mu} 89} \right)\end{matrix}$

In Expression 89, F_(R)(D_(r), ndm, noc) can be set in advance whenDoppler aliasing range D_(r), noc, and phase rotation amountsφ_(ndm)(noc) for applying Doppler shift amounts DOP₁, DOP₂, . . . , andDOP_(N_DM) are fixed. Therefore, for example, coded Dopplerdemultiplexer 212 may tabulate the correspondence between, on one hand,Doppler aliasing range D_(r), noc and the phase rotation amounts and, onthe other hand, F_(R)(D_(r), ndm, noc) and read F_(R)(D_(r), ndm, noc)based on Doppler aliasing range D_(r) and a phase rotation amount.

Since by the aliasing judgement processing it is possible to judge anindex (D_(rtrue)) that is a true Doppler aliasing range within theDoppler range of from 1/(2Loc×Tr) to less than 1/(2Loc×Tr) (in otherwords, it is possible to judge an index such that D_(rmin)=D_(rtrue)),it becomes possible for coded Doppler demultiplexer 212 to set, to zero,the correlation value between the orthogonal codes used for codemultiplexing in the Doppler range of from 1/(2Loc×Tr) to less than1/(2Loc×Tr), so as to perform the separation processing in which theinterference between the code multiplexed signals is suppressed.

Through the code separation processing as described above, and, based onthe aliasing judgement result assuming the Doppler range of±1/(2Loc×Tr), radar apparatus 10 can separate and receive the codedDoppler multiplexed signal to which DCI (ncm, ndm) used for themultiplexing transmission is assigned.

Further, since the coded Doppler multiplexed signal to which DCI (ncm,ndm) is assigned is transmitted from transmission antenna Tx #[ncm,ndm], it is also possible to judge transmission antenna 109. In otherwords, radar apparatus 10 can separate and receive the coded Dopplermultiplexed signal which is transmitted from transmission antenna Tx#[ncm, ndm] and to which DCI (ncm, ndm) is assigned.

In addition, for example, during coded Doppler demultiplexingprocessing, radar apparatus 10 performs, on the outputs of Doppleranalyzers 210 for each code element, Doppler phase correction, forexample, based on Doppler phase correction vector α(f_(s_comp_cfar),D_(r)) taking into consideration Doppler aliasing. Such phase correctioncorresponds to correcting phase changes corresponding to Dopplercomponents among the Doppler component candidates with respect tof_(s_comp_cfar). Mutual interference between code multiplexed signalscan thus be reduced, for example, as low as a noise level. In otherwords, radar apparatus 10 can reduce inter-code interference to suppressthe effect on degradation of the detection performance of radarapparatus 10.

The foregoing description has been given of an example of the operationof coded Doppler demultiplexer 212.

In FIG. 1, peak extractor 213 outputs, to direction estimator 214, atleast one of the outputs of Doppler analyzers 210 for distance indexf_(b_cfar) and Doppler frequency index f_(s_comp_cfar) inputted fromCFAR section 211. At this time, peak extractor 213 may use, for example,Dram′ that is a Doppler aliasing judgement result inputted from codedDoppler demultiplexer 212.

For example, in the example illustrated in FIG. 1, peak extractor 213outputs output VFT_(z) ¹(f_(b_cfar),f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin), ndm__(BF),1)/(N_(DM)+N_(int)(1)))) of first Doppler analyzer 210 (Doppler analyzer210-1)) to direction estimator 214. Here, ndm__(BF) is any one of 1, . .. , N_(DM), and a plurality of transmission antennas 109 to which thendm__(BF)-th Doppler multiplexed signal is assigned are a combination oftransmission antennas 109 that satisfies the condition of the adjacentarrangement described above, for example.

In FIG. 1, based on aliasing judgement result D_(rmin) for distanceindex f_(b_cfar) and Doppler frequency index f_(s_comp_cfar) inputtedfrom coded Doppler demultiplexer 212, direction estimator 214 performsdirection estimation processing for estimation of the direction of atarget based on separated received signal Y_(z)(f_(b_cfar),f_(s_comp_cfar), D_(rmin), ncm, ndm) of the coded Doppler multiplexedsignal to which DCI (ncm, ndm) is assigned and which is transmitted fromtransmission antenna Tx #[ncm, ndm], and based on the output from a partof Doppler analyzers 210 (Doppler analyzer 210-1 in FIG. 1) inputtedfrom peak extractor 213.

Note that, by way of example, the case where output VFT_(z)¹(f_(b_cfar), f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin), ndm__(BF),1)/(N_(DM)+N_(int)(1)))) from first Doppler analyzer 210 is used will bedescribed below, but the output from peak extractor 213 is not limitedto this. In addition, z=1, . . . , Na.

For example, direction estimator 214 generates, based on the outputs ofcoded Doppler demultiplexer 212 and peak extractor 213, virtualreception array correlation vector h(f_(b_cfar), f_(s_comp_cfar)) givenby following Expression 90 and performs the direction estimationprocessing.

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) includes Nt×Na elements, the number of which is theproduct of number Nt of transmission antennas and number Na of receptionantennas, and further includes elements resulting from use of beamtransmission antennas. Detailed descriptions will be given below.

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) includes elements of beam transmission antennas. Theelements of beam transmission antennas are based on the output (e.g.,VFT_(z) ¹(f_(b_cfar), f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin),ndm__(BF), 1)/(N_(DM)+N_(int)(1))))) of a part of Doppler analyzers 210that is inputted from peak extractor 213. The elements of beamtransmission antennas result from code multiplexing transmission isperformed using the same Doppler multiplexing and constitute a sub-arrayby adjacent transmission antennas 109 for orthogonal beam transmission.

For example, when there are N_(BF) beam transmission antennas, virtualreception array correlation vector h(f_(b_cfar), f_(s_comp_cfar))includes (Nt+N_(BF))×Na elements. By way of example, when number N_(BF)of beam transmission antennas is 1, virtual reception array correlationvector h(f_(b_cfar), f_(s_comp_cfar)) is expressed by followingExpression 90. In Expression 90, an example is expressed in which peakextractor 213 outputs output VFT_(z) ¹(f_(b_cfar),f_(s_comp_cfar)+(N_(code)F_(R)(D_(rmin), ndm__(BF))/N_(DM)+N_(int))))from first Doppler analyzer 210 to direction estimator 214, but thepresent invention is not limited to this.

Further, since the output of coded Doppler demultiplexer 212 and theoutput of peak extractor 213 have different noise levels, valuesobtained by multiplication by a normalizing factor may be used asvirtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)).

$\begin{matrix}\lbrack 100\rbrack & \; \\{{h\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}}} \right)} = \begin{Bmatrix}{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,1} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(1)}},1} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},1,N_{DM}} \right)} \\\vdots \\{Y_{1}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\{Y_{2}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\\vdots \\{Y_{Na}\left( {f_{{b\_}c{far}},f_{{s\_ comp}{\_ cfar}},D_{r\min},N_{{DOP\_ CODE}{(N_{DM})}},N_{DM}} \right)} \\{VF{T_{1}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{{nd}m_{BF}},1} \right)}}{N_{DM} + {N_{int}(1)}}}} \right)}} \\{{VF}{T_{2}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{{nd}m_{BF}},1} \right)}}{N_{DM} + {N_{int}(1)}}}} \right)}} \\\vdots \\{{VF}{T_{Na}^{1}\left( {f_{{b\_}c{far}},{f_{{s\_ comp}{\_ cfar}} + \frac{N_{code}{F_{R}\left( {D_{r\min},{{nd}m_{BF}},1} \right)}}{N_{DM} + {N_{int}(1)}}}} \right)}}\end{Bmatrix}} & \left( {{Expression}\mspace{14mu} 90} \right)\end{matrix}$

Virtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) is used in processing for performing, on reflected wavesignals from a target, direction estimation based on a phase differencebetween reception antennas 202.

Since subsequent operations are the same as those in Embodiment 1, thedescription thereof is omitted.

Through the above-described operations, Doppler shift setter 106 may setthe phase rotation amounts for applying Doppler shift amounts DOP₁,DOP₂, . . . , and DOP_(N_DM) variably for respective code elements ofthe orthogonal code sequences with code length Loc used for codemultiplexing. In other words, Doppler shift amounts DOP₁, DOP₂, . . . ,and DOP_(N_DM) may be set variably for respective code elements of theorthogonal code sequences. Even in this case, by the aliasing judgementprocessing of coded Doppler demultiplexer 212, it is possible to judgean index (D_(rtrue)) that is a true Doppler aliasing range within theDoppler range of from 1/(2Loc×Tr) to less than 1/(2Loc×Tr). Further, itbecomes possible for coded Doppler demultiplexer 212 to set, to zero,the correlation value between the orthogonal codes used for codemultiplexing in the Doppler range of from 1/(2Loc×Tr) to less than1/(2Loc×Tr), so as to perform the separation processing in which theinterference between the code multiplexed signals is suppressed.

In addition, as in Embodiment 1, when having N_(BF) beam transmissionantennas that perform orthogonal beam transmission by performing codemultiplexing transmission using the same Doppler multiplexing (e.g.,Doppler shift amount) and by forming a sub-array between adjacenttransmission antennas, radar apparatus 10 is capable of utilizing thetransmission antennas such that the number thereof is made greater thanthe number of transmission antennas used for multiplexing transmission.In this case, it is possible to include (Nt+N_(BF))×Na elements invirtual reception array correlation vector h(f_(b_cfar),f_(s_comp_cfar)). Accordingly, it is possible to improve the angularresolution or suppress the sidelobe level in radar apparatus 10.

The example of the operation in which the phase rotation amounts forapplying the Doppler shift amounts are variably set for respective codeelements of the orthogonal code sequences with code length Loc used forcode multiplexing has been described above.

Note that, here, the case of using the beam transmission antenna (e.g.,the case where Doppler shift amounts are the same between combinationsassociated respectively with adjacent transmission antennas of aplurality of transmission antennas 109 among a plurality of combinationsof Doppler shift amounts DOP_(ndm) and the orthogonal code sequences)has been described as an example, but the present disclosure is notlimited to this example. For example, when the Doppler shift amounts inthe combinations associated respectively with the adjacent transmissionantennas of a plurality of transmission antennas 109 differ from eachother, Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(N_DM) may beset variably for respective code elements of the orthogonal codesequences. Further, in the setting, among a plurality of combinations ofDoppler shift amounts DOP_(ndm) and the orthogonal code sequences, thenumber of multiplexing (number of coded Doppler multiplexing) by anorthogonal code sequence associated with at least one Doppler shiftamount DOP_(ndm) may differ from the numbers of coded Dopplermultiplexing associated with the other Doppler shift amounts (in otherwords, may be set non-uniformly). Alternatively, in the setting, among aplurality of combinations of Doppler shift amounts DOP_(ndm) and theorthogonal code sequences, the numbers of multiplexing by the orthogonalcode sequences (numbers of coded Doppler multiplexing) associated withDoppler shift amounts DOP_(ndm) may be the same (in other words, may beset uniformly).

(2) In a radar apparatus according to an exemplary embodiment of thepresent disclosure, a radar transmitter and a radar receiver may beindividually arranged in physically separate locations. In a radarreceiver according to an exemplary embodiment of the present disclosure,a direction estimator and any other component may be individuallyarranged in physically separate locations.

(3) The numeric values of parameters used in the exemplary embodiment ofthe present disclosure, such as number Nt of transmission antennas,number Na of reception antennas, number N_(DM) of Doppler multiplexing,number N_(CM) of codes, and number N_(BF) of beam transmission antennasare illustrative and are not limited to those values.

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes, for example, a central processing unit (CPU), astorage medium such as a read only memory (ROM) that stores a controlprogram, and a work memory such as a random access memory (RAM), whichare not illustrated. In this case, the functions of the sectionsdescribed above are implemented by the CPU executing the controlprogram. However, the hardware configuration of the radar apparatus isnot limited to that in this example. For example, the functionalsections of the radar apparatus may be implemented as an integratedcircuit (IC). Each functional section may be formed as an individualchip, or some or all of them may be formed into a single chip.

Various embodiments have been described with reference to the drawingshereinabove. Obviously, the present disclosure is not limited to theseexamples. Obviously, a person skilled in the art would arrive variationsand modification examples within a scope described in claims, and it isunderstood that these variations and modifications are within thetechnical scope of the present disclosure. Each constituent element ofthe above-mentioned embodiments may be combined optionally withoutdeparting from the spirit of the disclosure.

The expression “section” used in the above-described embodiments may bereplaced with another expression such as “circuit (circuitry),”“device,” “unit,” or “module.”

The above embodiments have been described with an example of aconfiguration using hardware, but the present disclosure can be realizedby software in cooperation with hardware.

Each functional block used in the description of each embodimentdescribed above is typically realized by an LSI, which is an integratedcircuit. The integrated circuit controls each functional block used inthe description of the above embodiments and may include an inputterminal and an output terminal. The LSI may be individually formed aschips, or one chip may be formed so as to include a part or all of thefunctional blocks. The LSI herein may be referred to as an IC, a systemLSI, a super LSI, or an ultra LSI depending on a difference in thedegree of integration.

However, the technique of implementing an integrated circuit is notlimited to the LSI and may be realized by using a dedicated circuit, ageneral-purpose processor, or a special-purpose processor. In addition,a Field Programmable Gate Array (FPGA) that can be programmed after themanufacture of the LSI or a reconfigurable processor in which theconnections and the settings of circuit cells disposed inside the LSIcan be reconfigured may be used.

If future integrated circuit technology replaces LSIs as a result of theadvancement of semiconductor technology or other derivative technology,the functional blocks could be integrated using the future integratedcircuit technology. Biotechnology can also be applied.

SUMMARY OF PRESENT DISCLOSURE

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes: a plurality of transmission antennas that transmita transmission signal; and a transmission circuit that applies a phaserotation amount corresponding to a Doppler shift amount and a codesequence to the transmission signal to perform multiplexing transmissionof the transmission signal from the plurality of transmission antennas,in which each of the plurality of transmission antennas is associatedwith a combination of the Doppler shift amount and the code sequencesuch that at least one of the Doppler shift amount and the code sequenceis different between a plurality of the combinations, and the Dopplershift amounts of those of the plurality of combinations which areassociated respectively with at least two adjacent transmission antennasof the plurality of transmission antennas are the same Doppler shiftamount, the at least two adjacent transmission antennas being a firstsub-array antenna.

In an exemplary embodiment of the present disclosure, a number ofmultiplexing by the code sequence associated with each of the Dopplershift amounts is the same among the plurality of combinations.

In an exemplary embodiment of the present disclosure, a number ofmultiplexing by the code sequence associated with at least one of theDoppler shift amounts is different from another number of multiplexingby the code sequence associated with another of the Doppler shiftamounts among the plurality of combinations.

In an exemplary embodiment of the present disclosure, the radarapparatus further includes: a plurality of reception antennas thatreceive a reflected wave signal that is the transmission signalreflected from a target; and a reception circuit that performs sensingprocessing for sensing the target using a virtual reception antennaconstituted by the plurality of transmission antennas, the plurality ofreception antennas, and the first sub-array antenna.

In an exemplary embodiment of the present disclosure, the transmissioncircuit controls directivity of the first sub-array antenna.

In an exemplary embodiment of the present disclosure, a transmissiontiming of the transmission signal is the same between the firstsub-array antenna and a second sub-array antenna constituted by at leasttwo adjacent transmission antennas of the plurality of transmissionantennas.

In an exemplary embodiment of the present disclosure, a transmissiontiming of the transmission signal is different between the firstsub-array antenna and a second sub-array antenna constituted by at leasttwo other adjacent transmission antennas of the plurality oftransmission antennas.

In an exemplary embodiment of the present disclosure, the same Dopplershift amount associated with a pair of the adjacent transmissionantennas differs for each transmission period in which a code element ofthe code sequence is transmitted.

In an exemplary embodiment of the present disclosure, the transmissionantenna has a sub-array configuration.

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes: a plurality of transmission antennas that transmita transmission signal; and a transmission circuit that applies a phaserotation amount corresponding to a Doppler shift amount and a codesequence to the transmission signal to perform multiplexing transmissionof the transmission signal from the plurality of transmission antennas,in which each of the plurality of transmission antennas is associatedwith a combination of the Doppler shift amount and the code sequencesuch that at least one of the Doppler shift amount and the code sequenceis different between a plurality of the combinations, and a number ofmultiplexing by the code sequence associated with each of the Dopplershift amounts is the same among the plurality of combinations.

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes: a plurality of transmission antennas that transmita transmission signal; and a transmission circuit that applies a phaserotation amount corresponding to a Doppler shift amount and a codesequence to the transmission signal to perform multiplexing transmissionof the transmission signal from the plurality of transmission antennas,in which each of the plurality of transmission antennas is associatedwith a combination of the Doppler shift amount and the code sequencesuch that at least one of the Doppler shift amount and the code sequenceis different between a plurality of the combinations, and those of theDoppler shift amounts the same as each other differ for eachtransmission period in which a code element of the code sequence istransmitted.

In an exemplary embodiment of the present disclosure, a number ofmultiplexing by the code sequence associated with each of the Dopplershift amounts is the same among the plurality of combinations.

In an exemplary embodiment of the present disclosure, a number ofmultiplexing by the code sequence associated with at least one of theDoppler shift amounts is different from another number of multiplexingby the code sequence associated with another of the Doppler shiftamounts among the plurality of combinations.

While various embodiments have been described herein above, it is to beappreciated that various changes in form and detail may be made withoutdeparting from the sprit and scope of the invention(s) presently orhereafter claimed.

This application is entitled to and claims the benefit of JapanesePatent Application No. 2020-204938, filed on Dec. 10, 2020, thedisclosure of which including the specification, drawings and abstractis incorporated herein by reference in its entirety.

INDUSTRIAL APPLICABILITY

The present disclosure is suitable as a radar apparatus for wide-anglerange sensing.

REFERENCE SIGNS LIST

-   10, 10 a, 20 Radar apparatus-   100, 300 Radar transmitter-   101 Radar transmission signal generator-   102 Transmission signal generation controller-   103 Modulation signal generator-   104 VCO-   105 Phase rotation amount setter-   106 Doppler shift setter-   107 Encoder-   108 Phase rotator-   109 Transmission antenna-   200 Radar receiver-   201 Antenna system processor-   202 Reception antenna-   203 Reception radio-   204 Mixer-   205 LPF-   206 Signal processor-   207 AD converter-   208 Beat frequency analyzer-   209 Output switch-   210 Doppler analyzer-   211 CFAR section-   212 Coded Doppler demultiplexer-   213, 213 a Peak extractor-   214 Direction estimator-   301 Directivity weight applier

1. A radar apparatus, comprising: a plurality of transmission antennasthat transmit a transmission signal; and a transmission circuit thatapplies a phase rotation amount corresponding to a Doppler shift amountand a code sequence to the transmission signal to perform multiplexingtransmission of the transmission signal from the plurality oftransmission antennas, wherein each of the plurality of transmissionantennas is associated with each of a plurality of combinations of theDoppler shift amount and the code sequence, each of the plurality ofcombinations is different at least one of the Doppler shift amount andthe code sequence, and the Doppler shift amounts of those of theplurality of combinations which are associated respectively with atleast two transmission antennas of the plurality of transmissionantennas are the same Doppler shift amount, the at least twotransmission antennas being a first sub-array antenna.
 2. The radarapparatus according to claim 1, wherein a number of multiplexing by thecode sequence associated with each of the Doppler shift amounts is thesame among the plurality of combinations.
 3. The radar apparatusaccording to claim 1, wherein a number of multiplexing by the codesequence associated with at least one of the Doppler shift amounts isdifferent from another number of multiplexing by the code sequenceassociated with another of the Doppler shift amounts among the pluralityof combinations.
 4. The radar apparatus according to claim 1, furthercomprising: a plurality of reception antennas that receive a reflectedwave signal that is the transmission signal reflected from a target; anda reception circuit that performs sensing processing for sensing thetarget using a virtual reception antenna constituted by the plurality oftransmission antennas, the plurality of reception antennas, and thefirst sub-array antenna.
 5. The radar apparatus according to claim 1,wherein the transmission circuit controls directivity of the firstsub-array antenna.
 6. The radar apparatus according to claim 1, whereina transmission timing of the transmission signal is the same between thefirst sub-array antenna and a second sub-array antenna constituted by atleast two transmission antennas of the plurality of transmissionantennas.
 7. The radar apparatus according to claim 1, wherein atransmission timing of the transmission signal is different between thefirst sub-array antenna and a second sub-array antenna constituted by atleast two other transmission antennas of the plurality of transmissionantennas.
 8. The radar apparatus according to claim 1, wherein the sameDoppler shift amount differs for each transmission period in which acode element of the code sequence is transmitted.
 9. The radar apparatusaccording to claim 1, wherein the transmission antenna has a sub-arrayconfiguration.
 10. A radar apparatus, comprising: a plurality oftransmission antennas that transmit a transmission signal; and atransmission circuit that applies a phase rotation amount correspondingto a Doppler shift amount and a code sequence to the transmission signalto perform multiplexing transmission of the transmission signal from theplurality of transmission antennas, wherein each of the plurality oftransmission antennas is associated with each of a plurality ofcombinations of the Doppler shift amount and the code sequence, each ofthe plurality of combinations is different at least one of the Dopplershift amount and the code sequence, and a number of multiplexing by thecode sequence associated with each of the Doppler shift amounts is thesame among the plurality of combinations.
 11. A radar apparatus,comprising: a plurality of transmission antennas that transmit atransmission signal; and a transmission circuit that applies a phaserotation amount corresponding to a Doppler shift amount and a codesequence to the transmission signal to perform multiplexing transmissionof the transmission signal from the plurality of transmission antennas,wherein each of the plurality of transmission antennas is associatedwith each of a plurality of combinations of the Doppler shift amount andthe code sequence, each of the plurality of combinations is different atleast one of the Doppler shift amount and the code sequence, and thoseof the Doppler shift amounts the same as each other differ for eachtransmission period in which a code element of the code sequence istransmitted.
 12. The radar apparatus according to claim 11, wherein anumber of multiplexing by the code sequence associated with each of theDoppler shift amounts is the same among the plurality of combinations.13. The radar apparatus according to claim 11, wherein a number ofmultiplexing by the code sequence associated with at least one of theDoppler shift amounts is different from another number of multiplexingby the code sequence associated with another of the Doppler shiftamounts among the plurality of combinations.